Open access peer-reviewed chapter

Power Converter Topologies for Multiphase Drive Applications

By Carlos A. Reusser

Submitted: August 16th 2018Reviewed: October 12th 2018Published: November 27th 2018

DOI: 10.5772/intechopen.81977

Downloaded: 214

Abstract

The yet growing demand for higher demanding industrial applications and the global concern about harmful emissions in the atmosphere have increased the interest for new developments in electric machines and power converters. To meet these new requirements, multiphase machines have become a very attractive solution, offering potential advantages over three-phase classical solutions. Multiphase machine’s power demand can be split over more than three phases, thus reducing the electric field stress on each winding (protecting the insulation system) and the requirements on maximum power ratings, for semiconductor devices. Moreover, only two degrees of freedom (i.e. two independently controllable currents) are required for independent flux and torque control. Due to the previous facts, the use of multiphase drives has become very attractive for applications and developments in areas such as electric ship propulsion, more-electric aircraft, electric and hybrid electric road vehicles, electric locomotive traction and in renewable electric energy generation. As a consequence of this multiphase drive tendency, the development of power converter topologies, capable of dealing with high power ratings and handling multiphase winding distributions, has encourage the development of new converter topologies, control strategies and mathematical tools, to face this new challenge.

Keywords

  • multiphase AC drive
  • neutral-point clamped
  • cascaded H-bridge
  • nine-switch converter
  • 11-switch converter
  • field-oriented control (FOC)
  • direct torque control (DTC)

1. Introduction

Multiphase variable-speed drives, based on multiphase AC machines, are nowadays the most natural solution for high-demanding industrial, traction and power generation and distribution applications [1].

The types of multiphase machines for variable-speed applications are in principle the same as their three-phase counterparts. These are synchronous machines, which depending on the excitation can be subclassified into wound rotor, permanent magnet or reluctance machines and induction machines.

In a multiphase winding, the stator winding distribution becomes more concentrated, rather than distributed, as in the case of three-phase windings. This fact and the particularity of using quasi-sinusoidal, rather than sinusoidal voltages because of the inverting process in the power conversion stage, have several advantages that can be summarized as lower field harmonics content and better fault tolerance, because of extra degrees of freedom and less susceptibility to torque pulsations, due to an excitation field with less harmonic content [2].

As a consequence, the use of a multiphase winding configuration improves the MMF spatial distribution, by reducing its harmonic content and losses, due to flux leakage, and increasing the machine’s efficiency. These facts have increased the interest for research and development for transportation applications, such as cargo ships, aircraft and road vehicles, thus also contributing to the reduction of greenhouse effect emissions.

2. Mathematical modeling of multiphase machines

All electrical machines can be found to be variations of a common set of fundamental principles, which apply alike the number of phases of which the machine is constructed.

In this context, multiphase machines can be treated as belonging to an n-dimensional space, corresponding to the respective state variables. The machine model, on its original phase-variable form, can be transformed into n/2two-dimensional subspaces, for a machine with an even number of phases. If the number of phases is odd, then the original n-dimensional subspace can be decomposed into a n1/2two-dimensional subspaces and a single-dimensional quantity, which corresponds to a common mode subspace.

Each new two-dimensional subspace is orthogonal to each other, so there is no mutual coupling and they are represented in a stationary reference frame, respecting all other state variables. These new two-dimensional subspaces are could be denoted as ukvk, where kstands for the respective new orthogonal subspace.

Let us consider an arbitrary state variable λ, defined as λ=λ1λ2λnT, and let Tbe a linear operator, which transforms the n-dimensional space into n/2two-dimensional subspaces, for neven, as in Eq. (1):

T=2n1cosαcos2αcos2αcosα0sinαsin2αsin2αsinα1cos2αcos4αcos4αcos2α0sin2αsin4αsin4αsin2α1cosn2αcos2n2αcos2n2αcosn2α0sinn2αsin2n2αsin2n2αsinn2αE1

In the case of nodd, then the corresponding transformation is given as in Eq. (2):

T=2n1cosαcos2αcos2αcosα0sinαsin2αsin2αsinα1cos2αcos4αcos4αcos2α0sin2αsin4αsin4αsin2α1cosn12αcos2n12αcos2n12αcos2n12α0sinn12αsin2n12αsin2n12αsin2n12α1212121212E2

The last row of Tin Eq. (2) corresponds to the projection of the state variable onto the common mode subspace.

Using the above-presented mathematical decomposition, any machine can be represented by an equivalent two-axis idealized machine, called the Kron’s primitive machine. The Kron’s primitive machine corresponds to the representation of the n-dimensional space, in the u1v1or αβsubspace, using the linear transformation Tnαβ; this definition can be found as the generalized theory of electrical machines.

The mathematical model describing a generalized n-dimensional electrical machine is given in Eqs. (3) and (4), for dynamics, respectively:

vs=Rsis+ddtψsE3
vr=Rrir+ddtψrE4
vs=vs1vs2vsnTis=is1is2isnTψs=ψs1ψs2ψsnTE5
vr=vr1vr2vrnTir=ir1ir2irnTψr=ψr1ψr2ψrnTE6

Considering symmetrical windings for both stator and rotor, then Rsand Rrare diagonal n×nmatrices, thus Rs=diagRsnand Rr=diagRrn. The stator and rotor flux linkages can be found as in Eqs. (7) and (8):

ψs=Lsis+LsrirE7
ψr=Lrir+LrsisE8

where due to the machine’s symmetry Lrs=LsrT; the corresponding stator and rotor inductance matrices are described as follows:

Ls=Ls11Ls12Ls1n1Ls1nLsn1Lsn2Lsnn1LsnnE9
Lr=Lr11Lr12Lr1n1Lr1nLrn1Lrn2Lrnn1LrnnE10

It has to be noticed that for both stator and rotor windings, L11=L22==Lnnwhich correspond to the winding self-inductance L¯; for the corresponding stator and rotor self-inductances within each one, due to the machine’s symmetry, it fulfills that Ljk=Lkj=Lmj,kjk. Then for the stator and rotor winding self-inductances, it can be stated that Ljj=Lσ+Lmwhere Lσcorresponds to the stator or rotor winding leakage inductance. On the other hand, the stator-to-rotor mutual inductance matrix Lsrcan be found as in Eq. (11):

Lsr=Mcosθrcosθrn1αcosθrn2αcosθrαcosθrαcosθrcosθrn1αcosθr2αcosθr2αcosθrαcosθrcosθr3αcosθrn1αcosθrn2αcosθrn3αcosθrE11

The resulting generalized machine representation is obtained by applying the linear transformation operator Tnαβto Eqs. (3) and (4), resulting in the following dynamic representation, in the αβsubspace, in Eqs. (12) and (13):

vsαβ=Rsisαβ+ddtψsαβE12
vrαβ=Rsirαβ+ddtψrαβjωrψrαβE13

The corresponding stator and rotor flux linkages given in Eqs. (7) and (8) are transformed into the αβsubspace as in Eqs. (14) and (15):

ψsαβ=Lsψsαβ+LsrirαβE14
ψrαβ=Lrirαβ+LrsisαβE15

The presented methodology considers a n-phase winding with uniform distribution that is α=2πn; in the case of machine with multiple groups of three-phase windings, αis the shifting angle between each group of three-phase windings. It has to be noted that for each three-phase winding transformation, an additional row has to be included in Eq. (2).

The developed electromechanical torque can be expressed in terms of the electromechanical energy conversion as in Eqs. (16) and (17):

Te=θrWfldisαβψsαβθrE16
Te=12iαβθrLiαβTE17
iαβ=isαβirαβTE18
L=LsLsrLsrTLrE19

Following the electromechanical torque can be expressed as in Eq. (20):

Te=12isαβθrLsrirαβTE20

The generalized machine representation given by Eqs. (12), (13) and (20) can be expressed in an arbitrary orthogonal synchronous reference frame dq, which rotates at synchronous speed ωk; thus the corresponding rotation into the dqstate variables is given by Eq. (21):

θk=ωkdtE21

Rotation into the dqsubspace is given by the unitary rotation matrix Udefined in Eq. (22):

U=cosθksinθksinθkcosθkE22

The generalized machine dynamic representation in the dqreference frame, as consequence of the rotation of Eqs. (12), (13) and (20), is given in Eqs. (23)(25), representing the Kron’s primitive machine model in a generalized synchronous reference frame:

vsdq=Rsisdq+ddtψsdq+jωkψsdqE23
vrdq=Rsirdq+ddtψrdq+jωkωrψrdqE24
Te=pLmLrψrdisqψrqisdE25

Equations (23) and (24) can be written in their matrix form as in Eq. (26), becoming a generalized impedance model, as described in Eq. (27):

vsdvsqvrdvrq=Rs+LsddtωkLsLmddtωkLmωkLsRs+LsddtωkLmLmddtLmddtωkωrLmRr+LrddtωkωrLrωkωrLmLmddtωkωrLrRr+LrddtisdisqirdirqE26
vdq=zidqE27

2.1 Multiphase synchronous machines

The use of multiphase synchronous machines has been focused on its application for medium- and high-power generation systems, being their primary use, on wind energy conversion systems (WECS). The previous statement is based on the fact that most of wind energy conversion systems operate in the low-voltage range, principally due to restrictions of winding insulation. This fact has stimulated the development of multiphase generator topologies, thus gaining increasing interest in the research for new converter topologies [3, 4]. Some of their main advantages are the following:

  • The total power can be divided into lower power converters.

  • Due to that each converter is insulated from each other, there is no circulating current between converters, which leads to no power derating for the converters.

  • The increase of the number of phases in the generator voltages is phase-shifted so that low-order harmonics are reduced and consequently smaller filters can be used.

  • Reduced torque pulsations.

  • Fault-tolerant redundancy under winding fault conditions.

  • Additional degrees of freedom which can be used to improve the machine performance.

Arrangement of multiple three-phase windings has become a very popular construction technique, for multiphase machines. In this field, the most common configuration is the six-phase machine, based on two independent three-phase windings, which are spatially shifted in 30 degrees, as shown in Figure 1.

Figure 1.

Dual three-phase stator winding configuration.

The use of multiple three-phase windings has the advantage of guaranteeing full decoupling under faulty conditions, thus preventing the circulation of common mode currents and pulsating torque.

The linear transformation operator Tis defined as in Eq. (28), with φ=π6:

T=131cos4φcos8φcosφcos5φcos9φ0sin4φsin8φsinφsin5φsin9φ1cos8φcos4φcos5φcosφcos9φ0sin8φsin4φsin5φsinφsin9φ111000000111E28

The equivalent model in the dqsynchronous subspace can be derived from Eq. (26); in this case it has to be noted that the synchronous reference frame dqcoincides with the rotor natural reference frame, thus ωk=ωr, becoming Eq. (29) and (30) for the stator:

vsd=Rsisd+ddtψsdωkψsqE29
vsq=Rsisq+ddtψsq+ωkψsdE30

Equations (31) and (32) correspond to the damping winding effect:

0=Rrird+ddtψrdE31
0=Rrirq+ddtψrqE32

The corresponding flux linkages, considering the general case of an anisotropic machine, are given in the following equations:

ψsd=Lsdisd+Lmdird+ψmE33
ψsq=Lsqisq+LmqirqE34
ψrd=Lrdird+Lmdisd+ψmE35
ψrq=Lrqirq+LmqisqE36

and the electromechanical torque is given in Eq. (37):

Te=pψmisq+LmdirdisqLmqirqisd+pLmdLmqisqisdE37

2.2 Multiphase induction machines

Modern industrial high-demanding processes are commonly based on induction machines. They are very attractive for these kind of applications because of their simplicity and capability to work under extreme torque demanding conditions [5].

Within the previous context, multiphase induction machines have become very popular for applications where high redundancy and power density are required. In particular, the use of multiple three-phase windings (six and nine phases) in naval propulsion systems has aroused much interest and encouraged the research of new multiphase converter topologies and control schemes [6, 7].

Based on the Kron’s primitive machine model developed in Eq. (26), the dynamical model of the multiphase induction machine is given in Eqs. (38)(41):

vsd=Rsisd+ddtψsdωkψsqE38
vsq=Rsisq+ddtψsq+ωkψsdE39
0=Rrird+ddtψrdE40
0=Rrirq+ddtψrqE41

Due to the isotropy of the induction machine and the absence of an MMF source in the rotor (permanent magnets or field winding), the corresponding flux linkages are given in Eqs. (42)(45):

ψsd=Lsisd+LmirdE42
ψsq=Lsisq+LmirqE43
ψrd=Lrird+LmisdE44
ψrq=Lrirq+LmisqE45

The electromechanical torque expression can be derived from Eq. (37) by considering ψm=0,Lmd=Lmq=Lm, becoming Eq. (46):

Te=pLmirdisqirqisdE46

3. Multiphase power converters

As stated previously, multiphase machines have many advantages over traditional three-phase-based machine drives, by reducing the impact of low-frequency torque pulsations and the dc-link current harmonic content. They also, due to the nature of their winding configuration, improve the system reliability, by introducing redundant operation conditions. As a consequence the use of multiphase converter topologies with multiphase drive arrangements has been proved as a viable approach for its application in high-demanding industrial applications.

In this field, the development of power converters capable to deal with the multiphase machine structure has capture much attention in the recent years; thus several topologies have been introduced in the last decades. This topologies can consist of arrangements of conventional two-level three-phase voltage source converters (2LVSC), multilevel converters (MLVSC) or in more specialized and dedicated topologies such as multiphase matrix converters.

3.1 Classical topologies

Classical topologies for multiphase converters are commonly based in arrangements of parallel-connected fundamental cells (multiple legs), or in multiple channel configuration, of voltage source converter topologies.

Commonly used topologies for multiphase applications are H-bridge converter (HBC), neutral-point clamped converter (NPC) and the cascaded H-bridge (CHB) topology, which are shown in Figure 2.

Figure 2.

Classical VSC topologies: (a) two-level bridge, (b) neutral-point clamped and (c) cascaded H-bridge.

Classical multiphase VSC topologies consist of an arrangement of nindividual half-cells connected in parallel to a single dc-link, as shown in Figure 3.

Figure 3.

Multiphase multicell topology.

Modulation of each individual half-cell circuit is implemented in such a way, to obtain nvoltage output signals shifted by 2π/n. Generation of the required voltage signals is supported by classical carrier-based strategies, such as sinusoidal pulse width modulation (SPWM) and space vector pulse width modulation (SVPWM).

SPWM methods can be implemented for two-level or multilevel half-cells. In the first case, only one high-frequency carrier is required for the respective switching signals per phase Sx, as given in Eqs. (46) and (48):

Sx=1uxNuc0uxN<ucE47
uxN=vxN12minvaNvbNvnN+maxvaNvbNvnNE48

where xstands for the corresponding phase x=1n, uxNis the reference to be synthesized for the corresponding phase, and ucstands for the carrier wave.

For multilevel parallel arrangements (NPC or CHB), the use of multiple carries, as an extension of the two-level PWM methods, has been proven as a suitable solution. Level-shifted PWM (LSPWM) has become a very popular modulation technique, because it fits for any multilevel converter topology and ensures low harmonic distortion. The corresponding switching states are given in Eq. (49):

Sxm=1uxNucm0uxN<ucmE49

where xstands for the corresponding phase x=1n, mto the corresponding level and, ucmis the carrier for the matched mlevel.

The main advantages of this topology are its simplicity by using half-bridges for each leg (phase), and the requirement of a single dc-link the feed the n-phase inverting stage. The use of a parallel arrangement of fundamental half-cells generates n1dimensional spaces, which can be decomposed into n1/2subspaces, ensuring redundant switching states.

Multicell topology also enables the possibility to have common or split connected loads. In the first case, each phase in the load side is connected to a common neutral point Nas presented in Figure 4; the main drawback with this configuration is the circulation of zero sequence currents, since the neutral points Nnare not insulated, thus establishing a common mode voltage vNn0in the dqsynchronous reference frame.

Figure 4.

Common connected load arrangement.

Split connected load arrangement is possible, if n=3kk=12, so the nphase system can be divided into kthree-phase insulated independent subsystems, as shown in Figure 5, with symmetric or asymmetric electrical shifting δ. It has to be noted that independently on the symmetric or asymmetric configuration, the electrical shifting between each phase on a single three-phase group is still 2π/3, constituting each individual symmetrical three-phase system.

Figure 5.

Split connected load arrangements. (a) Single-channel topology and (b) multichannel topology.

Figure 6a shows a split-phase single-channel arrangement where the corresponding phase for each half-cell is derived as y,y+k,y+2kwith =1k. A multichannel arrangement is shown in Figure 6b, which introduces some advantages, like full load dynamic decoupling and lower dc-link power rating. On the other hand, additional dc-link capacitors are required.

Figure 6.

mxn matrix converter equivalent model.

For both previously described topologies, SPWM is achieved by implementing the modulation as given in Eqs. (47) and (48) considering the asymmetry (if required) φfor each reference as uxNφ.

In the case of space vector modulation for multiphase converters, it is necessary to extend the space vector representation into its corresponding subspaces. So as presented previously in this chapter, the output voltage space vector can be decomposed into n/2orthogonal subspaces for n even and into n1/2orthogonal subspaces and single-dimensional quantity (common mode component) for nodd. Under this formulation, the reference voltage space vector to be synthesized can be expressed as in Eqs. (50) and (51) for neven and odd, respectively:

ux=u1u2unTE50
ux=u1u2un1TE51

For each orthogonal subspace, there exits two active vectors, v¯ukand v¯vk, corresponding to korthogonal subspace. Each active vector is applied for a duty-cycle δto synthesize the corresponding reference space vector, as in Eqs. (52) and (53):

ux=δu1v¯u1+δv1v¯v2+δu2v¯u2+δv2v¯v2++δun/2v¯un/2+δvn/2v¯vn/2E52
ux=j=1n/2δujv¯uj+j=1n/2δvjv¯vjE53

In the case of nodd, an additional degree of freedom is introduced by the common mode voltage vcmwhich represents a single quantity and not a space vector as in Eq. (54):

ux=j=1n1/2δujv¯uj+j=1n1/2δvjv¯vj+δcmvcmE54

Eqs (53) and (54) can be interpreted as the partitioning of the αβstationary reference frame, into n!or n1!adjacent sectors, and an additional common mode component, orthogonal to the αβsubspace. Sector identification can then be achieved by implementing Eq. (55) where Scorresponds to the given sector within the αβsubspace:

S=θM!+1M=neven number of subspaces.n1oddnumber of subspaces.E55

3.2 Matrix converter

The matrix converter is a direct AC-AC converter, which uses an arrangement of bidirectional switches, to connect each input phase, with a single corresponding output phase, thus generating an arrangement of mxnpower switches, where mis the number of input phases and nthe number of output phases. In Figure 6, the equivalent model of a mxnmatrix converter is shown.

Due to the absence of a dc-link stage, the output voltages should be synthesized by selecting segments of the input voltages, by generating the adequate switching states. However, some switching state restrictions have to be taken into account, because of the particular topology of the matrix converter. Let’s consider a generalized switching state Sjk, such that

Sjk=1switchSjkis inonstate.0switchSjkis inoffstate.j=1,2,,mk=1,2,,nE56

For every instant t, the switching state Sjkmust comply with both conditions stated in Eqs (57) and (58), thus meaning that for all tonly one input phase is connected to one output phase, avoiding short-circuit condition, and also all output phases are connected to at least one input phase, ensuring no open-circuit condition. This last condition is intended to protect the bidirectional switches that cannot handle reverse current flow, due to the inductive energy discharge:

j=1nSjk=1E57
Sjk=1k=1,2,,nE58

Sinusoidal PWM scheme is implemented via the Venturini method [8], which is based on the solution of the relational input-output equations of the matrix converter, given in Eqs. (59)(61):

Vo=MViM=δ11δ12δ1mδ21δ22δ2mδn1δn2δnmE59
Ii=MTIoE60
Pi=PoE61

where Voand Vistand for the output and input voltage space vectors, respectively, Ioand Iifor the output and input current space vectors, Mis the low-frequency matrix or modulation index transfer matrix, and Poand Picorrespond to the output and input active power.

On the other hand, space vector PWM formulation has no difference, as it is implemented in VSC. However, its complexity lies in the fact in that the absence of a nearly constant dc-link voltage, from which the reference voltage space vectors are synthesized. So both space vectors are to be composed using the input and output, voltage and current space vectors simultaneously. Moreover, the total number of possible switching states is 2nm, from which the forbidden conditions have to be considered. Space vector implementation is largely explained in literature [9].

The matrix converter has several advantages over multiphase voltage source converters, for multiphase drive applications. It is capable to synthesize nearly sinusoidal output voltage and currents, with low-order harmonics, thus improving the MMF distribution in machine air gap, eliminating torque ripple and preventing mechanical stresses on the output shaft. It provides bidirectional energy flow and provides full power factor control. Moreover, due to the absence of dc-link, it presents more power density, because of the lack of large capacitors, becoming an alternative for integrated drive converter configurations [10, 11]. However the complexity in the implementation of SPWM and SVPWM schemes, and the complex commutation strategy for the bidirectional switches, makes the matrix converter less attractive than the voltage source-based multiphase solutions.

There exist several variations of the direct matrix converter topology presented previously, such as the indirect and sparse matrix converter topologies, which are extensively described in literature [12]. These topological variations relay on the same basis but differ in the number and type of power switches used.

3.3 Nine-switch converter

The topology is derived from two three-phase voltage source converters that share a positive and a negative busbar, respectively, as shown in Figure 7.

Figure 7.

(a) Classical arrangement with 12 switches. (b) Nine-switch converter topology.

The nine-switch converter has the ability to operate in back-to-back mode, as rectifier (A stage input) and inverter (B stage output), as two channel rectifier (A and B stages input) and as two stage inverter (A and B stages output), enabling the converter to handle a six-phase systems with just one channel, in spite of commonly 12-switch back-to-back topologies.

However, this topological advantage introduces some drawbacks to the nine-switch topology, because of the fact that Sx, Syand Szswitches are shared by both converter stages A and B. Hence 3 switches are eliminated of the 12 needed for multiphase operation as in multichannel topology, some forbidden switching states are introduced, thus remaining 27 allowed states, and also the maximum output voltage gain is limited [13]. In Figure 8, the allowed switching states per leg are shown.

Figure 8.

Nine-switch converter allowed switching states.

Carrier-based PWM modulation schemes such as sinusoidal PWM (SPWM), space vector modulation (SVM) and min-max (third harmonic injection) PWM [14] can be applied to the nine-switch converter, using two independent SPWM modulation schemes, one for each converter stage, as in the 12-switch back-to-back converter. However, because of the restrictions introduced to the modulation, pattern by the middle switches S4, S5and S6, its switching pattern can be obtained as in Eq. (62):

Sk+3=SkSk+6¯k=1,,3E62

The implementation of SPWM considers the use of two voltage reference space vectors xand y, for each three-phase winding group. The NSC can operate either in constant (CFM) or variable frequency (VFM) modes, depending on each space vector angular frequency. However, due to the nature of the particular application in multiphase drives, the NSC is to be operated in the constant frequency mode, as presented in Figure 9.

Figure 9.

CFM modulation.

For the constant frequency mode operation (CFM), both reference space vectors xand yhave the same angular frequency ωk, preserving its spatial shifting δ, so they are to be found at least in contiguous sectors of the voltage hexagon, as shown in Figure 10a, being the maximum voltage gain ggiven in Eq. (63):

g=13δ2+π3E63

Figure 10.

Space vector decomposition. (a) Single carrier, (b) shifted multi-carrier and (c) line-line voltage.

By introducing a phase shifting of π/6between both PWM carriers, a second group of active vectors is introduced as shown in Figure 10b, which results in a voltage gain as in Eq. (64):

g=13δ+π6E64

Recently, the NSC has been gaining much attention in various applications like isolated wind-hydro hybrid power system, power quality enhancement and hybrid electric vehicles because of its ability to interconnect multiphase power systems in common or split configurations and also independent three-phase-based systems, in constant or variable frequency modes. Figure 11 shows simulation results for a wind energy conversion system (WECS) based on a nine-switch back-to-back converter topology [15].

Figure 11.

Simulation results for a back-to-back NSC for wind energy conversion.

Another important feature of the NSC is the capability to rearrange its switching states under single- or multiple-switch faulty conditions [16], as shown in Figure 12.

Figure 12.

Nine-switch converter reconfiguration under faulty operation.

3.4 11-switch converter

The 11-switch converter, presented in [17], consists of a modified topology of the 9-switch converter topology, previously discussed. As shown in Figure 13, this topology introduces two additional switches Sx,Sy, whose main propose is to mitigate the common mode voltage during the zero switching states.

Figure 13.

11-switch converter.

4. Control of multiphase electric drives

As discussed previously in this chapter, the electromechanical torque developed by the multiphase machine depends only on the state variables in αβsubspace, thus meaning that the other subspaces do not contribute to the energy conversion process but only losses. This fact makes possible the implementation of oriented control schemes, such as field-oriented control (FOC), direct torque control (DTC) and model-based predictive control (MBPC), in a synchronous reference frame dq, by rotating the state variables in αβsubspace.

An additional control loop in the dqreference frame can be implemented, to compensate the losses in the remaining subspaces. Control goals for the multiphase drive can be summarized as maximum torque per Ampere operation, control of nominal flux and control of rotor speed/torque.

4.1 Field-oriented control

The characteristics of field-oriented control (FOC) have made this control strategy the most widely used for high-demanding industrial applications. Field-oriented control is based in the decoupling of the current space vector, into a flux-producing component and a torque-producing component. This is achieved by rotating the current space vector from αβsubspace, into the synchronous rotating reference frame dq, oriented with respect to the rotor flux linkage space vector, as shown in Figure 14.

Figure 14.

dq synchronous reference frame rotation.

In this way the magnetizing flux can be controlled, so that the machine operates with nominal flux under any condition, and also torque can be controlled only by isq, because ψrq=0, due to the orientation of ψrwhich only exists in the direct axis orientation. Implementation of the FOC scheme for a multiphase AC drive is shown in Figure 15.

Figure 15.

Field-oriented control scheme of a dual three-phase winding induction machine.

Figure 16 shows simulation results for a dual three-phase induction machine drive using FOC scheme, under load impact.

Figure 16.

Dual three-phase winding induction machine FOC scheme simulation results.

The main drawbacks of FOC are the requirement in the estimation of the rotor flux linkage space vector and the rotation of the state space variables into this synchronous reference frame, becoming a very complex process. Also, as the dynamic response, this control strategy is limited by maximum bandwidth achievable for the PI controllers, which represents one of the fundamental limitations of linear controllers.

4.2 Direct torque control

Direct torque control (DTC) is based on the estimation of torque and flux directly from the state variables of the AC machine. The torque and flux can be controlled by applying the suitable voltage vector, synthesized by the available switching states of the converter.

The required voltage vector is chosen via a switching table, as function of the actuation of the torque and flux loop hysteresis controllers (in terms of increasing or decreasing flux or torque for a certain operational point). Implementation of the DTC scheme for a multiphase AC drive is shown in Figure 17.

Figure 17.

Direct torque control scheme of a dual three-phase winding PM synchronous machine.

The main characteristics of DTC are its simple implementation and a fast dynamic response achieved by using hysteresis controllers. Furthermore, the required switching states are directly assigned from the switching table algorithm, so no modulator is needed.

5. Conclusions

In this chapter, the main advantages of multiphase machine drives for its application in high-demanding industrial processes, traction and renewable energy grid interfacing were presented, with their main focus on multiphase power converter topologies. Various technical issues related to classical multiphase converters, based on multicell arrangements, were discussed, as well as some new converter topologies, such as the nine-switch converter (NSC) and 11-switch converter (ESC), were introduced.

The main advantages of classical multiphase converter topologies, based on voltage source converters (VSC), are mostly referred to their topological simplicity and their capability to implement conventional sinusoidal-PWM-based techniques. On the other hand, the increasing number of semi-conductors and dc-link capacitors (in the case of multichannel arrangements) and the need for common mode current compensation are their major drawbacks. In this field of application, matrix converters arise as suitable alternative, which enables the possibility to handle multiple output phases. However, maximum voltage gain limitations and the complexity of the modulation and commutation strategies are their main disadvantages, when compared to classical topologies.

Nine-switch and eleven-switch converters appear as a middle-point alternative between multicell and matrix converter topologies. These topologies allow the use of sinusoidal-PWM-based modulation techniques, without the need of complex modulation and commutation strategies (as in the case of the matrix converter), using a single dc-link stage.

Implementation of control strategies for multiphase drive, such as field-oriented control (FOC) and direct torque control (DTC), has been easily achieved by using multi-space decomposition of the each state space vectors (representing a state space variable).

© 2018 The Author(s). Licensee IntechOpen. This chapter is distributed under the terms of the Creative Commons Attribution 3.0 License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

How to cite and reference

Link to this chapter Copy to clipboard

Cite this chapter Copy to clipboard

Carlos A. Reusser (November 27th 2018). Power Converter Topologies for Multiphase Drive Applications, Electric Power Conversion, Marian Găiceanu, IntechOpen, DOI: 10.5772/intechopen.81977. Available from:

chapter statistics

214total chapter downloads

More statistics for editors and authors

Login to your personal dashboard for more detailed statistics on your publications.

Access personal reporting

Related Content

This Book

Next chapter

Five-Phase Line Commutated Rectifiers

By Mahmoud I. Masoud

Related Book

First chapter

Synthesis Processes for Li-Ion Battery Electrodes – From Solid State Reaction to Solvothermal Self-Assembly Methods

By Verónica Palomares and Teófilo Rojo

We are IntechOpen, the world's leading publisher of Open Access books. Built by scientists, for scientists. Our readership spans scientists, professors, researchers, librarians, and students, as well as business professionals. We share our knowledge and peer-reveiwed research papers with libraries, scientific and engineering societies, and also work with corporate R&D departments and government entities.

More About Us