New Electronic Devices for Power Converters

Abstract

Power electronic devices are crucial components of power converter systems. The evolution of power devices drives the development of power converters, including improvements in performance, reliability, and power capacity. In this chapter, the author expounds the structure, working principle, and static and dynamic characteristics of the conventional PN junction diode. And the silicon carbide (SiC) Schottky barrier diode (SBD), junction barrier Schottky (JBS) diode, trench JBS (T-JBS) diode, and sidewall-enhanced trench JBS (SET-JBS) diode are also discussed and compared. Also, the structures and properties of the gallium oxide (Ga2O3) SBD and heterojunction diode are also summarized. Next, the author gives a detailed analysis and discussion of the silicon power metal-oxide-semiconductor field-effect transistor (MOSFET), superjunction MOSFET, and the SiC MOSFET and JFET, and the Ga2O3 MOSFET. Then, the device structure and operating principle, switching characteristics, and current tailing mechanism of the insulated gate bipolar transistor (IGBT) are also analyzed and summarized in detail. Finally, the energy band structure, working principle, and switching characteristic of the gallium nitride (GaN) high-electron mobility transistor (HEMT), one of the hot devices in the current market, are also described. Finally, the summary and prospect of power electronic devices are also presented in this chapter.

1. Introduction

Power electronic devices are the core components of power converters and directly affect the performance and reliability of power converters. In recent years, in addition to conventional silicon-based devices, some new electronic devices have emerged, which are widely used in power conversion systems and play an important role in the performance improvement and development of power converters. This chapter explains the traditional silicon-based power diodes and power MOSFET and also describes the structures and working principles of superjunction power MOSFET devices and IGBT power devices. More importantly, with the development of the wide-bandgap semiconductor technology, SiC diodes, SiC MOSFETs, SiC JFETs, and GaN HEMTs are also widely used in various power converters and power electronic systems, so this chapter also describes the wide-bandgap power semiconductor devices. At the same time, the ultra-wide bandgap power semiconductor devices represented by gallium oxide (Ga₂O₃) have also become a research hotspot, and this chapter also explains the Ga₂O₃ power diodes and Ga₂O₃ power MOSFETs. Finally, the development trend of new electronic devices is also summarized.

2. Power diodes

2.1 Silicon PN junction diode

Silicon power diodes are the most commonly used power electronic devices, and their basic principle is the unidirectional conductivity of the PN junction diode. When a P-type doping and a N-type doping are performed on a semiconductor material, a PN junction is formed at the interface. Due to the existence of the doping concentration gradient at the interface, the holes in the P-type region diffuse to the N-type region and recombine with the majority carrier electrons in the N-type region. Similarly, the electrons in the N-type region diffuse to the side of the P-type region and recombine with the majority carrier holes of the P-type region. And a space charge region is formed at the interface, in which the N-type side has only positive charges ionized by the donors, and the P-type region has only negative charges ionized by the acceptors. Therefore, an electric field directed from the N-type region to the P-type region is formed at the space charge region. And under the action of this electric field, the minority carrier holes in the N-type region drift toward the P-type region, and the minority carrier electrons in the P-type region also drift toward the N-type region. The diffusion and drift motions of carriers will eventually reach a dynamic equilibrium, and the width of the space charge region (depletion region) remains constant, and the built-in electric field (E) is also maintained constant. Figure 1 shows the equilibrium PN Junction and its space charge region.

Figure 1.

Equilibrium PN junction and its space charge region.

The diode is formed when the PN junction chip is packaged and the anode (A) and cathode (K) electrodes are led out. When the PN junction (or diode) is forward biased, since the external electric field (E_V) and the built-in electric field (E) are in opposite directions, the space charge region is narrowed, and the diffusion effect of the majority carriers is greatly enhanced at this time, forming a larger forward current (I_F), the PN junction is turned on, which is shown in Figure 2.

Figure 2.

PN junction or diode in the forward bias conduction state.

On the contrary, when the PN junction (or diode) is reverse biased, since the external electric field (E_V) and the built-in electric field (E) are in the same direction, the space charge region becomes wider, which greatly hinders the diffusion of majority carriers. And the PN junction is in the off state, and only a negligible reverse leakage current (I_R) flows through the diode, as shown in Figure 3. Figure 3 also shows the electric field distribution of the reverse-biased diode. When the applied reverse bias voltage (V) increases, the peak electric field (E_max) also increases accordingly. When the E_max is up to the critical breakdown electric field (E_C) of the semiconductor, the diode breaks down, the applied voltage is the breakdown voltage (U_BR), which is also equal to the area of the electric field distribution triangle. When the diode breaks down, the reverse current of the diode will increase sharply.

Figure 3.

PN junction diode in the reverse bias state.

The I-V characteristic curve of the PN junction diode is shown in Figure 4. When the forward voltage drop (V_AK) is higher than the turn-on (or knee) voltage drop (V_ON) the diode is turned ON, and the current (I_AK) is approximately exponential with respect to the voltage (V_AK). When the diode is reverse biased, its reverse leakage current (I_R) is very small and can be negligible, but when the reverse bias voltage crosses to U_BR, the current increases sharply, so the U_BR is the breakdown voltage of the diode. When designing a power converter, it is necessary to be reasonable in choosing the U_BR and forward current capability of the diode. The I-V characteristic of the PN diode can be described as Eq. (1).

Figure 4.

The I-V characteristics of the PN junction diode.

$${\mathit{I}}_{\mathbf{AK}}={\mathit{I}}_{\mathit{S}}·\left[{\mathbf{e}}^{\left({\mathit{qV}}_{\mathbf{AK}}/\mathit{k}\mathbf{T}\right)}-\mathbf{1}\right]$$E1

Where I_S is the reverse state saturation current (leakage current) I_R, T is the thermodynamic temperature, k is the Boltzmann constant, and V_AK is the applied bias voltage between the anode and the cathode. At room temperature, the kT/q is about 26 mV.

Due to the charge storage effect in the diode, the switching of the diode from the ON state to the OFF state requires a transient process. Figure 5 shows the reverse recovery transient of the diode. When the power supply voltage of the diode circuit changes from the forward state to the reverse state of the diode, the current of the diode decreases from the forward on-state current (I_F) to the reverse saturation current I_R, it does not maintain at I_R immediately, but increases reversely to I_RM, which is because although the voltage of the external circuit has been reversed, the inside of the diode is still full of carriers, and these carriers need a process to be extracted from the body of the diode. This period of time is called the reverse recovery time (t_rr) of the diode, and the charge extracted during the t_rr time is called the reverse recovery charge (Q_rr). Generally speaking, the t_rr and Q_rr of PN junction diodes with the same rated voltage and rated current are larger than those of Schottky diodes.

Figure 5.

The reverse recovery characteristics of the PN junction diode.

2.2 SiC power diodes

The energy bandgap of 4H-silicon carbide (4H-SiC) is about 3 times that of Si (Silicon), the thermal conductivity is also 3 times that of Si, the critical breakdown electric field is about 8 to 10 times that of Si, and the saturation drift velocity of electrons is 2 times that of Si. These superior properties of SiC make it the preferred material for high-frequency, high-power, high-temperature, and radiation-resistant semiconductor devices.

Figure 6 shows the specific on-resistance (R_on,sp) of N-type drift region in 4H-SiC and Silicon at different breakdown voltages. And the R_on,sp of the 4H-SiC drift region is about 2000 times smaller than that of the silicon devices for the same breakdown voltage [1].

Figure 6.

Specific on-resistance of n-type drift region in 4H-SiC and silicon at at different breakdown voltages.

Since the turn-on voltage (or knee voltage) of the SiC PN junction is as high as about 2.8 V, which is much higher than that of the SiC Schottky diode with a value of lower than 1 V. So, the commercial SiC diodes with a breakdown voltage of less than 4500 V are almost Schottky diodes. For the SiC Schottky diodes, the much lower drift region resistance and much higher energy bandgap compared with silicon can boost the breakdown voltage to over 3000 V with reasonable on-state voltage drop (V_F) and relatively low leakage current [1]. Although silicon-based Schottky diodes are also commonly used in power systems and power converters, they operate at low voltages (typically ≤200 V). They offer very low on-state voltage drops and losses despite high leakage current and low maximum operating temperature.

The SiC power diode structures are mainly Schottky barrier diodes (SBD) and junction barrier Schottky diodes (JBS), which are shown in Figure 7a, b, respectively. The main feature of the SiC SBD is the Schottky contact formed at the interface between the metal and 4H-SiC. While the SiC JBS introduces P-type regions at a certain distance in the SBD to shield the electric field at the Schottky contact interface and reduce the reverse leakage current. The energy band diagram for the metal–semiconductor (4H-SiC) contact is shown in Figure 7c.

Figure 7.

(a) SiC Schottky barrier diodes (SBD), (b) SiC junction barrier Schottky diodes (JBS) and (c) the band diagram for metal–semiconductor contact.

In the typical normal operating state of the above two devices, the on-state current is dominated by majority carriers—electrons, and the storage effect of the minority carriers in the drift region is almost negligible. This causes the transition from the on-state to the reverse blocking state of the SiC SBD and JBS diode very fast with a much shorter reverse recovery time (t_rr) and a much lower reverse recovery charge (Q_rr) compared with those of the silicon PN diode. The high switching speed and the high current density compared with the silicon PN diodes make them suitable for high frequency, high power, and high-end applications.

For SiC Schottky power diodes, due to the low doping concentration in the N-type drift (N-drift) region to support high reverse blocking voltages, the current via thermionic emission current transport mechanism is dominant in Schottky barrier diodes. So the thermionic emission theory can be used to describe the current density J_AK flows across the Schottky barrier interface [2], which is shown in Eq. (2):

$$J_{\mathrm{AK}}=\mathrm{A}*{\mathrm{T}}^2{\mathrm{e}}^{-\left(\mathrm{q}{\mathrm{\Phi }}_{\mathrm{BN}}/\mathrm{kT}\right)}\left[{\mathrm{e}}^{\left({\mathit{qV}}_{\mathrm{AK}}/\mathrm{kT}\right)}-1\right]$$E2

where A* is the effective Richardson constant, ${\Phi }_{\mathrm{BN}}$ is the barrier height of the metal–semiconductor contact (shown in Figure 7c), T is the thermodynamic temperature, k is the Boltzmann constant, and V_AK is the applied bias voltage between the anode and the cathode. Among them, the Richardson constant of N-type silicon carbide material is 146 A·cm⁻²·K⁻².

The turn-on voltage drop (V_ON) of the Schottky diode is mainly determined by the barrier height (${\Phi }_{\mathrm{BN}})$. And the ${\Phi }_{\mathrm{BN}}$ mainly depends on the workfunction of metal materials, thus the on-state voltage drops of SiC Schottky diode can be selected by choosing different metal materials.

Based on the operation mechanism of SiC Schottky diode, the static I-V characteristic curves of SiC SBD and JBS diodes are shown in Figure 8. As can be seen from the figure, the SiC SBD exhibits a higher forward current density, but it also shows a larger leakage current and a lower breakdown voltage (U_BR). Although the SiC JBS diode has a lower current density, the leakage current in the blocking state is much lower and the U_BR is also higher than those of the SBD. This is because the P-type regions introduced in the JBS structure reduce the Schottky contact area, resulting in a certain reduction in current density, but the introduction of the P-type regions shields the electric field at the Schottky contact interface, thereby effectively reduces the reverse leakage current and improves the breakdown voltage. However, in SiC materials, the ion implantation depth of the P-type regions is relatively shallow (usually <1 μm), thus the electric field shielding effect is limited. Then, the trench JBS (T-JBS) diode has been proposed to achieve a low reverse leakage current [3, 4]. Unfortunately, the T-JBS structure introduces a severe JFET (junction field-effect transistor) effect, which greatly reduces the on-state current density. And the sidewall-enhanced JBS (SET-JBS) diode was proposed to alleviate the JFET effect and increase the Schottky contact area, resulting in an improvement in current density [5]. Figure 9a, b shows the device structures of the T-JBS diode and the SET-JBS diode, respectively. And the static I-V characteristic comparison result is also shown in Figure 9c.

Figure 8.

The static characteristic curves of SiC SBD and JBS diodes.

Figure 9.

(a) SiC trench JBS (T-JBS) diode, (b) SiC sidewall enhanced JBS diode (SET-JBS), and (c) static characteristic comparison of different Schottky diodes [5].

2.3 Ga₂O₃ power diodes

Gallium oxide (Ga₂O₃) is a representative material of the ultra-wide bandgap semiconductor material and has attracted extensive research interest in recent years. There are five isomers of Ga₂O₃, and the beta-Ga₂O₃ (β-Ga₂O₃) is mostly used material for power devices. Due to its ultra-wideband gap (over 4 times of Si), high theoretical breakdown electric field (8MV/cm), large Baliga figure of merit (3400), and stable chemical properties, it has become an ideal choice for high-voltage and high-power rectifiers and field-effect transistors [6, 7]. Nowadays, with the improvement of crystal growth technology, large-scale Ga₂O₃ single crystals have been produced, such as pulling method, guided mode method, and floating zone melting method [8]. There is also a method of heteroepitaxial growth of gallium oxide thin films on substrates such as quartz glass, sapphire, silicon, and gallium arsenide [9, 10, 11].

Figure 10a shows the cross-sectional structure view of the Ga₂O₃ Schottky barrier diode (SBD). As shown in the figure, the top metal layer (Pt/Ti/Au) is in contact with the Ga₂O₃ N-type drift region to form a Schottky contact. And the bottom metal (Ti/Au) is in contact with a heavily doped N+ Ga₂O₃ substrate to form an ohmic contact. The operating mechanism of Ga₂O₃ SBD is similar to that of SiC SBD. However, compared with SiC materials, the effective P-type doping has not yet been achieved in Ga₂O₃ materials. Due to the lack of P-type doping in the Ga₂O₃ materials, Figure 10b shows a structure of a Ga₂O₃ heterojunction (HJ) PN diode [12]. The current density of the Ga₂O₃ heterojunction PN diode is higher than that of the Ga₂O₃ SBD due to the hole injection and conductance modulation effect in the N-type drift region. However, because the barrier height of the heterojunction PN junction is higher than that of Schottky contact, the turn-on voltage drop of the HJ PN diode is higher than that of the SBD, but the Ga₂O₃ heterojunction PN diode has great advantages in the field of ultra-high voltage (e.g., > 6500 V) applications with a lower voltage drop at the same current density compared with Ga₂O₃ SBD. At present, the Ga₂O₃ power heterojunction diode with a breakdown voltage exceeding 8000 V has been developed [12].

Figure 10.

Device structures of (a) Ga₂O₃ SBD, (b) Ga₂O₃ heterojunction PN diode [9].

Figure 11 shows the forward I-V characteristic curves of the Ga₂O₃ SBD and the Ga₂O₃ HJ PN diode. And it can be seen from the figure that the turn-on voltage drop of the Ga₂O₃ SBD (V_ON1) is lower than that of the Ga₂O₃ HJ PN diode (V_ON2), but the current I_AK of the Ga₂O₃ HJ PN diode rises faster with the voltage V_AK, which is mainly caused by the conductance modulation effect in the Ga₂O₃ HJ PN diode.

Figure 11.

The Ga₂O₃ SBD and Ga₂O₃ HJ PN diode forward I-V characteristics.

2.4 Dynamic characteristic comparison of the diodes

Figure 12 presents an intuitive rough comparison result of the reverse recovery characteristics of Si, SiC, and Ga₂O₃ diodes at the same rated breakdown voltage and rated current. It can be seen from the figure that the Si PN junction diode has the largest reverse recovery current, the longest reverse recovery time, and the largest reverse recovery charge. And the Ga₂O₃ SBD has the best reverse recovery performance. The reverse recovery performance of the SiC Schottky diodes is between that of Si and Ga₂O₃ diodes.

Figure 12.

Comparison of reverse recovery characteristics of Si, SiC, and Ga₂O₃ diodes at the same rated breakdown voltage and rated current.

3. Power MOSFET and JFET

3.1 Device structure and operating mechanism

The power MOSFET (metal-oxide-semiconductor field-effect transistor) has many advantages: (a) it is a voltage-controlled device with high input impedance and low driving power consumption, (b) it is no secondary breakdown with wide safe operating area (SOA), and due to good thermal stability, the operating temperature can reach up to 200°C, which is 50°C higher than that of the bipolar transistor (BJT), (c) it is a majority carrier conduction device with the strong anti-irradiation ability and (d) it is no minority-carrier storage effect, and the switching frequency is high. Because the power MOSFET has those advantages mentioned above, it has always been a research hotspot in the industry and widely used in the power converters.

Figure 13a, b shows the device structure of the N-channel vertical power MOSFET and its electric field distribution in the blocking state, respectively. When the gate (G) to source (S) voltage (V_GS) is higher than the threshold voltage (V_TH) of the N-channel MOSFET and the drain-to-source voltage (V_DS) is positive, the MOSFET is turned ON, and the electrons flow from the source n + region through the channel and n-drift region to the n + drain region to form the drain current I_DS. Conversely, if V_GS < V_TH and V_DS is a positive high voltage, the channel is off and the MOSFET is in a forward blocking state (voltage sustaining state). At this time, the electric field distribution inside the device is roughly as shown in Figure 13b, and the voltage is mainly sustained by the n-drift region. Figure 13c, d demonstrates the symbols of the N-channel power MOSFET and P-channel power MOSFET, respectively.

Figure 13.

(a) Device structure of the N-channel power MOSFET, (b) the electric field distribution of the power MOSFET in the blocking state; the symbols of the (c) N-channel power MOSFET and (d) P-channel power MOSFET.

3.1.1 Static characteristic

Figure 14a, b shows the transfer characteristic and the output characteristic of the N-channel power MOSFET, respectively. And when V_GS > V_TH, the MOSFET is turned ON, and the drain current I_DS increases as the V_GS increases. From the output characteristic curve, when the device is in the cut-off region, I_DS is almost negligible; when the device is in the triode region, the current I_DS increases sharply with V_DS. While in the saturation region, I_DS hardly increases with V_DS. And when the device is in the breakdown region, the current I_DS increases sharply, causing a dramatic increment in power consumption, which may cause a thermal runaway of the device. Therefore, the device should be avoided as much as possible to operate in the breakdown region.

Figure 14.

(a) The transfer characteristic curve and (b) output characteristic curve of the power MOSFET.

3.1.2 Dynamic characteristic

Figure 15a, b shows the power MOSFET switching test circuit with an inductive load L and the typical characteristics of power MOSFET switching transients [13]. As the power MOSFET is a majority carrier device, there is no minority carrier storage effect, so the switching speed is fast, typically 20–50 ns. As shown in Figure 15b, the turning-on time of the device is t_sw(on) (t_sw(on) = t_d1 + t_on), and the turn-off time is t_sw(off) (t_sw(off) = t_d2 + t_off). Typically, most devices have a longer turn-off time than their turn-on time.

Figure 15.

(a) MOSFET switching test circuit with inductive load, (b) typical characteristics of power MOSFET switching transients [13].

3.2 Superjunction power MOSFET

The conventional power MOSFET devices have an inherent contradiction that the specific on-resistance (R_on,sp) is proportional to the 2.5th power of the breakdown voltage (BV), that is R_on,sp∝ BV^2.5, which is dreaded as “silicon limit” theory. This means that even with a small increase in BV, R_on,sp will increase substantially, thereby greatly increasing the conduction loss of the device.

In 1993, Chen invented the superjunction device, which greatly improved the contradiction between R_on,sp and BV with a much better relationship as R_on,sp∝ BV^1.32 [14, 15]. And the superjunction MOSFET was commercialized in 1998 and hailed as a “milestone” in the field of power electronic devices [16].

Figure 16a, b illustrate the device structure of superjunction MOSFET and its approximate electric field distribution in the drift region. Due to the introduction of the charge compensation effect of the P-type pillars in the drift region, the doping concentration of the n-drift can be greatly increased, thereby greatly reducing the R_on,sp. of the device. At the same time, the charge compensation effect makes the total net charge of the properly designed superjunction drift region to be zero in the blocking state, so that the electric field distribution is approximately rectangular. Therefore, at the same rated voltage, the drift region thickness of superjunction MOSFETs is thinner than that of conventional power MOSFETs with a triangular electric field distribution. The much higher n-drift doping concentration and thinner n-drift region thickness greatly reduce the R_on,sp. of the device, enabling superjunction devices to break the “silicon limit” of conventional MOSFETs.

Figure 16.

(a) Structure of superjunction MOSFET and (b) the approximate electric field distribution in the drift region.

Figure 17 shows the R_on,sp. comparison result between the silicon conventional power MOSFET and superjunction MOSFET at different breakdown voltages [17]. As can be seen from the figure, the relationship between the R_on,sp. and BV of the superjunction MOSFET is approximately linear. The comparison results show that the superjunction devices can be used for higher power density and higher-end applications.

Figure 17.

R_on,sp comparison between the silicon conventional MOSFET and Superjunction MOSFET at different breakdown voltages [17].

3.3 SiC power MOSFET

Figure 18 presents the structural comparison of Si and SiC vertical power MOSFETs at the same rated breakdown voltage. It can be seen from the figure that under the same rated breakdown voltage, the thickness of the n-drift region of the SiC MOSFET is about 1/10 of that of the silicon MOSFET, so the drift region resistance is dramatically reduced and the current density of the SiC power MOSFET is greatly improved, the conduction loss is greatly reduced, the switching speed is also improved, and the chip size is greatly reduced. At the same time, due to the larger energy band gap and higher thermal conductivity of SiC materials, the SiC MOSFETs can be operated at temperatures over 200°C. However, compared with silicon MOSFETs, the channel mobility of SiC MOSFETs is still very low, and its on-state resistance still has a large room for improvement. At the same time, the electric field of the gate oxide layer may be very high, which brings challenges to the reliability of the gate oxide layer.

Figure 18.

Structural comparison of Si and SiC vertical power MOSFETs at the same breakdown voltage.

Since the current capability of SiC devices is much larger than that of silicon devices, SiC lateral devices can also be used in power-integrated circuits to handle larger power conversions. And the SiC lateral device can easily achieve over 1200 V breakdown voltage while still obtaining a low on-resistance [18, 19]. Figure 19 shows the cross-sectional view of a SiC lateral power MOSFET device [20]. Compared with the SiC vertical MOSFET, all electrodes of the SiC lateral power MOSFET are on the surface of the device, so that the SiC lateral MOSFET can be integrated with SiC low-voltage integrated circuits on the same chip to realize monolithic SiC power integrated circuits. The operating principle of the SiC lateral power MOSFET is almost the same as the vertical power MOSFET. The only difference between the two kinds of devices is that the current of the lateral power MOSFET flows laterally, and the electron flow path is shown by the red dotted line in the figure. It is worth mentioning that the purpose of introducing the p-top region into the lateral power MOSFET of this device is to increase the doping concentration of n-drift through the principle of charge compensation, and greatly reduce the on-resistance while optimizing the surface electric field. So the device has the advantages of high breakdown voltage and low on-resistance.

Figure 19.

Cross-sectional view of a SiC lateral power MOSFET device [18].

3.4 SiC JFET

Due to the extremely low channel mobility of SiC MOSFETs and the reliability issues of the SiO₂ gate oxide layer, SiC JFET (junction field-effect transistor) devices were once favored by researchers and the industry, and have also been commercialized [21, 22]. Figure 20a, b show the structure and the symbol of the SiC power JFET device, respectively. Different from the MOSFET, the SiC JFET controls the turn-off of the device by applying a negative voltage through the p⁺ gate with respect to the source to completely deplete the N-type channel region. Generally, the SiC JFET is a normally on device, and when the gate-to-source is zero biased (V_GS = 0 V), the channel is not fully depleted, the device is in the on-state, and electrons flow from the n⁺ source through the channel region to the n⁺ drain. Since the channel mobility of SiC JFETs is much greater than that of SiC MOSFETs, the SiC JFETs have lower on-resistances. However, the gate control voltages of the two devices are different. The V_GS of the JFET cannot be higher than the turn-on voltage drop of the gate-source PN junction (∼ 2.8 V), while the V_GS of the MOSFET can be as high as 20 V.

Figure 20.

(a) The cross-sectional view and (b) symbol of the SiC power JFET; (c) SiC JFET/Si MOSFET cascode configuration; (d) SiC SBD-JFET.

In order to take the performance advantages of SiC JFETs and make SiC JFETs as easy to control as MOSFETs at the same time, a cascode configuration consisting of a low-voltage high-current Si MOSFET and a high-voltage SiC JFET has emerged on the market and gained lots of applications, which is shown in Figure 20c [23]. In addition, in order to realize the self-reverse recovery of the SiC JFET device and improve its performance, a new SiC SBD-JFET has been proposed in Figure 20d [24].

3.5 Ga₂O₃ power MOSFET

Figure 21a shows the structure of the depletion-mode Ga₂O₃ MOSFET, which has a negative threshold voltage V_TH [25]. The two heavily Si-doped N⁺ regions are connected with metals to form low-resistance ohmic contacts, respectively. The source-connected field plate above the gate can effectively reduce the surface electric field and improve the breakdown voltage of the device in the blocking state. And due to the existence of the Fe-doped semi-insulated β-Ga₂O₃ substrate, the leakage current through the substrate is obviously reduced. During forward conduction, the gate-to-source voltage V_GS is higher than V_TH, the channel region under the gate is not fully depleted, and a positive drain voltage relative to the source V_DS is applied, the electrons flow from the N⁺ source to the drain along the channel region and the N-type β-Ga₂O₃ drift region to form the on-state drain current I_DS. When the gate-to-source voltage V_GS < V_TH, the channel region is fully depleted, and the device is changed to the blocking state with no electron flowing from the source to drain. And the maximum blocking voltage (breakdown voltage) is mainly determined by the distance from the gate to the drain, the doping concentration of N-type β-Ga₂O₃, and the parameters of the field plate. Figure 21b shows a vertical enhancement-mode Ga₂O₃ MOSFET with a breakdown voltage over 1 kV [26].

Figure 21.

(a) Depletion-mode Ga₂O₃ lateral MOSFET [25], (b) enhancement-mode Ga₂O₃ vertical transistor [26].

The transfer characteristic curve of the Ga₂O₃ MOSFET is shown in Figure 22a. It can be seen from the figure that the V_TH of the depletion-mode device is negative, and the V_TH of the enhancement-mode device is positive. Since there is no effective P-type doping in Ga₂O₃, therefore, the common devices are almost in depletion mode. The main way of realizing enhancement-mode Ga₂O₃ MOSFET is to make the channel region to be very thin (such as the recessed gate structure [27]) or to be very narrow (as shown in Figure 21b), so that when the V_GS is zero biased, the channel region can also be completely depleted. Although the realization of enhancement-mode devices increases the complexity of the fabrication process, the enhancement-mode devices are easier to be controlled from the application point of view. Figure 22b plots the output I-V curves of the Ga₂O₃ MOSFET and both the depletion-mode and enhancement-mode Ga₂O₃ MOSFETs have similar I-V curves. Also, it can be seen that the transfer characteristics and I-V characteristics of the Ga₂O₃ MOSFET are similar to those of the Si and SiC power MOSFETs described above, but due to the different material parameters, the current capability of the Ga₂O₃ MOSFET device is higher. In addition, the switching characteristics of Ga₂O₃ MOSFETs are similar to those of Si and SiC MOSFETs.

Figure 22.

(a) Transfer characteristic curve of the depletion-mode and enhancement-mode Ga₂O₃ MOSFETs (b) I-V output characteristic curve of Ga₂O₃ MOSFETs.

4. Insulated gate bipolar transistor (IGBT)

Insulated gate bipolar transistor (IGBT) is a composite fully controlled voltage-driven power electronic device composed of BJT (bipolar junction transistor) and MOSFET, which has both the high input impedance and the low on-state voltage drop. It was once hailed as an almost ideal switching device in the field of power electronics, except for its higher turn-off loss and longer turn-off time compared with those of the power MOSFETs.

Figure 23a, b demonstrate cross-section view of the IGBT and the equivalent circuit of internal structure, respectively [28]. From the perspective of device structure, the n⁺ region connected to the emitter (E) electrode, p-base, gate, and n-drift region constitute an N-channel MOSFET (N-MOSFET). At the same time, the p⁺/p-base, n-drift/n-buffer regions, and the p⁺ collector (C) constitute a PNP BJT. Therefore, from the perspective of the internal device structure, the IGBT can be regarded as a combination of an N-MOSFET and a PNP BJT. Thus, the IGBT has the advantages of the high input resistance of N-MOSFET and the large current density of BJT. It is worth noting that many researchers also call the electrode collector (C) on the back of the device as anode (A), and the electrode emitter (E) on the surface as cathode (K). Figure 23c also shows the symbol of the IGBT.

Figure 23.

(a) Cross-section view of the IGBT, (b) equivalent circuit of internal structure, (c) the symbol of the IGBT.

In the blocking state, the gate voltage with respect to the emitter (V_GE) is zero biased or negatively biased. At this time, the N-MOSFET controlled by the gate is in the off-state, and the IGBT is also in the off-state. And the applied positive voltage between the collector electrode and the emitter electrode (V_CE) is sustained by the P-base/N-drift junction. Since the doping concentration of the p-base region is much higher than that of the n-drift region, so the breakdown voltage of the IGBT is mainly determined by the thickness and doping concentration of the n-drift region.

In the on-state, V_GE is applied to a positive voltage (usually +15 V). And an inversion layer electron channel connecting the n⁺ region and the n-drift region is formed on the surface of the p-base region under the gate. The electrons flow from the n⁺ region through the channel into the n-drift region and finally into the p⁺ collector region. The electron current acts as the base drive current of the PNP transistor, which facilitates the injection of holes from the p ⁺ collector region into the n-buffer and n-drift regions, and finally into the emitter electrode.

Figure 24a, b illustrate the transfer characteristic and output characteristic curves of the IGBT, respectively. As can be seen from the figure, the transfer characteristic curve of the IGBT is similar to that of power MOSFET. However, the forward output characteristic of the IGBTs is slightly different from that of MOSFETs. The main difference is that, in addition to V_GE > V_TH, the forward conduction of IGBTs requires V_CE to be higher than the turn-on voltage V_PN of the p⁺/n-buffer PN junction. After the turning-on of the device, a large number of holes are injected into the n-drift region from the p⁺ collector, resulting in a conductance modulation effect, which greatly increases the current density and reduces the on-state voltage drop (V_ON) of the device. Figure 24b also reveals the reverse breakdown characteristic of the IGBT, the reverse breakdown voltage is mainly determined by the breakdown voltage of the p⁺/n-buffer junction, usually because the doping concentration of the n-buffer region is much higher than the n-drift region, the reverse breakdown voltage of the IGBT is very low. However, for an IGBT without an n-buffer region, the reverse breakdown voltage may also be close to the forward breakdown voltage.

Figure 24.

(a) Transfer characteristic and (b) output characteristic curves of the IGBT.

The turning-on characteristic of the IGBT is similar to that of the power MOSFET, but its turning-off characteristic is different from that of the power MOSFET. Figure 25 shows the typical switching-off characteristic of the IGBT [29]. Compared with the switching-off transient of the power MOSFET, the turning-off transient of the IGBT is much longer and a long tail current is appeared in the turning-off process [30]. The reason is that after the channel of the IGBT is turned off, a large number of nonequilibrium electrons in the drift region flow out to the p⁺ collector region under the action of the electric field. During this transient, the bottom p⁺/n-buffer PN junction is still in the forward biased state, and holes are continuously injected into the n-drift region until almost all the electrons are extracted from the n-drift region. Thus, the holes injection during the turn-off process is the main reason for the current tailing. Although there are some design optimizations and structural improvements to increase the turn-off speed of IGBTs, most of them come at the expense of forward voltage drop (V_ON) [31]. And the improvements in the trade-off relationship between the turn-on voltage drop and turn-off loss of the IGBT are still being pursued [32]. Nevertheless, the switching speed of IGBT is far inferior to that of MOSFET, which limits the high-frequency application of the IGBT.

Figure 25.

Typical switching off characteristic of the IGBT [29].

5. GaN high electron mobility transistor (HEMT)

In 1992, the first Gallium nitride high electron mobility transistor (GaN HEMT) was developed by using metal organic chemical vapor deposition (MOCVD) [33]. Subsequently, GaN devices have attracted great attention and research. The GaN HEMT has the advantages of high current density, high breakdown voltage, high operating frequency, high reliability, and low switching loss. GaN HEMTs have great potential for application in high frequency, high efficiency, and high power density power electronic systems. Currently, the GaN HEMTs are commercialized with rated voltage up to 650 V and are widely used in power converters, power adapters, on-board charging, data centers, and other applications [34].

For nitride semiconductors, the thermodynamic stable phase is a hexagonal symmetric wurtzite structure, while the thermodynamic metastable phase is a cubic symmetric sphalerite structure [35]. The wurtzite GaN crystal structure does not show symmetry along the C-axis, besides the sum of the vector P of the polarization intensity of Ga-N covalent bond is not zero. There should be a deviation between positive and negative ions, so a strong spontaneous polarization effect generates inside the GaN. As for AlGaN/GaN heterojunctions, the lattice constants of the two materials do not match, leading to the existence of stress forces between atoms near the contact surface of the two materials. Under the action of this stress force, the lattice asymmetry is enhanced, meanwhile the lattice deformation deviates the center of positive and negative charges in the lattice, resulting in a strong piezoelectric polarization effect.

Figure 26a, b show the energy band diagram and the structure of the enhancement-mode GaN HEMT, respectively [36, 37]. The heterojunction energy is discontinuous due to the strong total polarized induced electric field generated by the addition of piezoelectric polarization and spontaneous polarization in the heterostructure, as well as the conduction band shift ΔE_c at the heterojunction interface. The energy band bends in the GaN layer to form a triangular electron potential well, which captures electrons ionized by donor impurities and then forms a large number of two-dimensional electron gas (2DEG). Due to the existence of a high potential barrier on the side of AlGaN, it is hard for electrons to cross the potential well, therefore, electrons are restricted to move laterally in the thin layer of the interface, instead of moving perpendicular to the interface. Different from the channel electrons in traditional MOSFET, 2DEG accumulates on one side of the intrinsic potential well layer, realizing the separation of carriers and the Coulomb scattering center. There is almost no electron impurity scattering in the potential well, which indicates that the 2DEG has a very high electron mobility. At the same time, the concentration of 2DEG obtained under the unintentional doping of GaN is quite considerable, thereby the GaN HEMT devices have high current density and unique application value.

Figure 26.

(a) GaN HEMT basic energy band diagram, (b) enhancement-mode GaN HEMT basic configuration [36, 37].

1. Static characteristic

   Figure 27a, b show the transfer characteristics and output characteristics of the enhancement-mode GaN HEMT, respectively. The threshold voltage (V_TH) of the GaN HEMT is the gate-to-source voltage (V_GS) corresponding to the device from off state to on state, that is, the V_TH is the voltage applied to the gate when the AlGaN and GaN interface forms the 2DEG. Currently, the threshold voltage (V_TH) of the typical commercial AlGaN/GaN HEMT is positive—enhancement mode HEMT.

   It is found that the 2DEG characteristics are very similar to the channel electrons of MOSFET, so the output characteristics of the enhanced GaN HEMT are very close to the characteristics of n-channel MOSFET. By varying the applied drain-source voltage when the device is turned on, GaN HEMT can be operated in the linear (unsaturated) and saturation regions. However, in the GaN HEMT, when V_DS is high, the drain current I_DS decreases with the increase of V_DS, and the current collapse effect occurs. The main reason is that under the large V_DS, a high electric field is generated between the gate and drain, and the channel hot electrons are excited to tunnel to the surface of AlGaN, and are trapped by the surface states between the gate and drain, forming a virtual gate and a current collapse phenomenon [38].

2. Dynamic characteristics

   High concentration of 2DEG with high mobility exists in the channel layer of the GaN HEMT, which on the one hand enables the device to form a maximum forward current, and on the other hand enables it to operate at high frequency and high power. Compared with traditional Si-based and SiC MOSFETs, the GaN HEMT has a faster switching time and smaller switching loss. Figure 28 shows the typical switching waveforms of an enhancement-mode GaN HEMT in a double-pulse test circuit [39]. As can be seen from the figure, the turn-on and turn-off times of the GaN HEMT are very short, both around 10 ns. However, it is worth noting that different GaN HEMT devices and applications in different circuits have slightly different switching times.

Figure 27.

(a) Enhancement mode GaN HEMT transfer characteristics curve, (b) GaN HEMT DC output characteristics curve.

Figure 28.

Switching characteristics of the enhancement-mode GaN HEMT: (a) turn-on, (b) turn-off (redrawn from figure 1 in ref. [39]).

6. Prospects for new electronic devices

For power electronic devices, we always pursue higher breakdown voltage, lower loss, higher reliability and thermal stability, and low cost. In recent years, with the increasing demand for electronic devices in power electronic systems and power converters, new electronic devices represented by SiC devices and GaN devices have also achieved rapid development. However, there is still much room for improvement in the performance of these devices, and these new devices will continue to achieve breakthroughs in performance and cost reductions in the future. At the same time, besides the Ga₂O₃ devices, new electronic devices based on ultra-wide bandgap materials (such as diamond, BN, and AlN) will also emerge one after another and will be gradually applied in the market for high voltage, high-power, and high-end applications. Figure 29 shows the relationship between specific on-resistance and breakdown voltage of power electronic devices based on various materials [40], which also shows the development trend of power device materials from another perspective.

Figure 29.

Relationship between specific on-resistance and breakdown voltage of power electronic devices based on various materials [40].

7. Conclusions

The emergence and development of new power electronic devices are critical to the development of power converters and power electronic systems. Understanding how electronic devices work is important for better design of power converters. This chapter describes in detail the power electronic devices commonly used in power converters. Starting from the structure and working principle of PN junction, this chapter describes the structure and main properties of SiC and Ga₂O₃ power diodes. And the structure and characteristics of power MOSFET, superjunction MOSFET, SiC MOSFET, SiC JFET, and Ga₂O₃ MOSFET are described. Then, the structure, principle, and characteristics of IGBT, an extremely important bipolar device in modern power electronic systems, are described. Finally, the structure, working principle and related characteristics of the emerging GaN HEMT devices are also described in detail. And looking ahead, new power electronic devices, such as diamond diodes and diamond MOSFETs, will continue to appear and develop to meet the more stringent requirements of power converters for lower loss, higher breakdown voltage, higher power density, higher switching frequency, and reliability.