\r\n\tThe book welcomes topics such as: basics of milling process, including traditional and modern methods of milling, milling parameters, mechanism and kinetics of milling etc, equipment used for milling, different types of materials fabricated by milling and their applications, milling processes (dry milling and wet milling), microstructure and properties of products are also presented, modeling and simulation of milling process, milling and equipment.
",isbn:null,printIsbn:"979-953-307-X-X",pdfIsbn:null,doi:null,price:0,priceEur:null,priceUsd:null,slug:null,numberOfPages:0,isOpenForSubmission:!1,hash:"cc2da5436da998d3f9ed1660ab80f9f5",bookSignature:"Dr. Mohsen Mhadhbi",publishedDate:null,coverURL:"//cdnintech.com/web/frontend/www/assets/cover.jpg",keywords:"Milling, Traditional Methods, Modern Methods, Mechanism and Kinetics, Equipment Models, Size Reduction, Media, Nanostructured Materials, Proprieties, Wet Milling, Dry Milling, Crystallinity,Theory, Milling Forces, Analytical Modelling, Applications",numberOfDownloads:null,numberOfWosCitations:0,numberOfCrossrefCitations:null,numberOfDimensionsCitations:null,numberOfTotalCitations:null,isAvailableForWebshopOrdering:!0,dateEndFirstStepPublish:"February 25th 2019",dateEndSecondStepPublish:"March 18th 2019",dateEndThirdStepPublish:"May 17th 2019",dateEndFourthStepPublish:"August 5th 2019",dateEndFifthStepPublish:"October 4th 2019",remainingDaysToSecondStep:"9 months",secondStepPassed:!0,currentStepOfPublishingProcess:5,editedByType:null,kuFlag:!1,editors:[{id:"228366",title:"Dr.",name:"Mohsen",middleName:null,surname:"Mhadhbi",slug:"mohsen-mhadhbi",fullName:"Mohsen Mhadhbi",profilePictureURL:"https://mts.intechopen.com/storage/users/228366/images/system/228366.jpeg",biography:"Dr. Mohsen Mhadhbi has Ph.D. in Chemistry from the Faculty of Sciences of Sfax at the University of Sfax (Tunisia). Since 2011, he is an Assistant Professor and member of a research team of the Laboratory of Useful Materials focusing on synthesis and characterization of nanomaterials for industrials responders (hydrogen storage, cutting tools and biomedical applications) at National Institute of Research and Physical-chemical Analysis (Tunisia). Mhadhbi has served as a teacher in Inorganic Chemistry in different Institutes in Tunisia. He has supervised several researchers in materials science. He published work in national and international impacted journals and books. Mhadhbi is a member of different associations. 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From chapter submission and review, to approval and revision, copyediting and design, until final publication, I work closely with authors and editors to ensure a simple and easy publishing process. I maintain constant and effective communication with authors, editors and reviewers, which allows for a level of personal support that enables contributors to fully commit and concentrate on the chapters they are writing, editing, or reviewing. I assist authors in the preparation of their full chapter submissions and track important deadlines and ensure they are met. I help to coordinate internal processes such as linguistic review, and monitor the technical aspects of the process. As an ASM I am also involved in the acquisition of editors. 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Chan and Manoj Kumar Tiwari",coverURL:"https://cdn.intechopen.com/books/images_new/3794.jpg",editedByType:"Edited by",editors:[{id:"252210",title:"Dr.",name:"Felix",surname:"Chan",slug:"felix-chan",fullName:"Felix Chan"}],productType:{id:"1",chapterContentType:"chapter",authoredCaption:"Edited by"}},{type:"book",id:"3621",title:"Silver Nanoparticles",subtitle:null,isOpenForSubmission:!1,hash:null,slug:"silver-nanoparticles",bookSignature:"David Pozo Perez",coverURL:"https://cdn.intechopen.com/books/images_new/3621.jpg",editedByType:"Edited by",editors:[{id:"6667",title:"Dr.",name:"David",surname:"Pozo",slug:"david-pozo",fullName:"David Pozo"}],productType:{id:"1",chapterContentType:"chapter",authoredCaption:"Edited by"}}]},chapter:{item:{type:"chapter",id:"47290",title:"Antennas for Body Centric Wireless Communications at Millimeter Wave Frequencies",doi:"10.5772/58816",slug:"antennas-for-body-centric-wireless-communications-at-millimeter-wave-frequencies",body:'
1. Introduction
Over the past few years, wearable antennas for body centric wireless communication systems have been increasingly gaining attention. Body area networks (BANs) are wireless communication systems that enable communications between wearable and/or implanted electronic devices.
Such systems are of great interest for various applications such as healthcare, entertainment, military, identification systems, sport, smart home, and space [1, 2]. Indeed, portable devices, incorporating antennas close to the human body, have been used for many decades by the military. Nowadays, in order to increase overall effectiveness of soldiers on the battlefield, works are undertaken to integrate wireless systems to all equipment, such as weapons, sighting systems, helmet, and so on. Besides, wearable or implanted sensors increase the ability of doctors to monitor their patients at long distance and in real time. This monitoring capability is also used in sport applications and in rescue worker interventions.
Recently, studies carried out at millimeter waves have grown rapidly. This is due to the fact that many advantages can be found in operating such systems at millimeter waves compared to microwaves. First, because of the large available spectrum (7 GHz worldwide), very high data rates can be reached (up to 5 Gb/s) [3]. Second, it provides a high level of security and low interference with adjacent networks [4]. Finally, compared to on-body devices operating at microwaves, the size of similar millimeter-wave systems is significantly reduced.
Hence, the use of millimeter wave systems for BANs will have a high impact, in particular in the defense sector (Fig. 1), where communications emanating from a dismounted soldier leads to detection, location and vulnerability to enemy attack. The high atmospheric attenuation in the 60-GHz band will lead to much higher levels of security against detection, interception and jamming. Fig. 1 illustrates a scenario of soldier-to-soldier communications for covert battlefield operation where co-located soldiers are wirelessly networked to allow high-speed communications within a cluttered urban warfare environment. Besides, every soldier is equipped with advanced technology significantly improving situational awareness, lethality and survivability such as GPS, helmet mounted display, RADAR bullet detector, etc.
In addition, millimeter wave BANs will also benefit civilian sectors such as healthcare, personal entertainment, sports training, and emergency services. In hospital, clinics, entertainment venues, and public transport, there is a need to relay personalized data to and from individuals, in confined areas, or in crowds, and the high frequency and highly directive beams from small millimeter-wave antennas will reduce interference between users and other communication equipment.
The aim of this book chapter is to provide a review of recent progresses and outstanding challenges in the field of antennas for body-centric communication at millimeter waves.
Figure 1.
Soldier-to-soldier communications for covert battlefield operations. The black arrows represent some possible wireless links allowing data transfer from one soldier to another.
2. Electromagnetic properties and modeling of the human body
In this Section, the electromagnetic properties and modeling of the human body are investigated. First, to study the interaction of millimeter waves and the human body, the skin dielectric properties are carefully characterized. Then, the influence of the antenna feeding is investigated. Then, as the dielectric properties of the skin have been assessed, a numerical model of the human body is introduced using a Debye model. Finally, a semi-solid phantom is introduced for antenna measurement in close proximity to the body.
2.1. Interaction of millimeter waves with the human body
The primary biological targets of 60-GHz radiations are the skin and eyes. Exposure of the eyes leads to the absorption of the EM energy by the cornea characterized by a free water content of 75% and a thickness of 0.5mm. Ocular lesions have been found after high-intensity exposure of the eye (3W/cm2, 6min) [5]. However, studies performed at 60 GHz (10mW/cm2, 8h) demonstrated no detectable physiological modifications [6], indicating that millimeter waves act on the cornea in a dose-dependent manner.
Hereafter we will essentially consider the interactions with the skin as it covers 95% of the human body surface. From the EM viewpoint, human skin can be considered as an anisotropic multilayer dispersive structure made of three different layers, namely, epidermis, dermis, and subcutaneous fat layer (Fig. 2). The skin also contains capillaries and nerve endings. It is mainly composed of 65.3% of free water, 24.6% of proteins, and 9.4% of lipids [7].
Figure 2.
Schematic representation of the skin structure.
Knowledge of the dielectric properties of the skin is essential for the determination of the reflection from, transmission through, and absorption in the body, as well as for EM modeling. In contrast to frequencies below 20 GHz, existing data on the permittivity of tissues in the millimeter-wave band are very limited [8]-[11] due to some technical difficulties. In the 10–100 GHz range, the dispersive dielectric properties of the skin and biological solutions are primarily related to the rotational dispersion of free water molecules. In particular, high losses are related to the free water relaxation with the peak at 23 GHz at 33°C.
In contrast to frequencies below 20 GHz, the already-existing data on the relative permittivity of human tissues at millimeter waves are very limited. In addition, the results reported so far in the literature strongly depend on the measurement technique, the sample type (in vivo or in vitro study) and other experimental conditions such as skin temperature, location on the body and thickness of different skin layers.
Table 1 provides a summary of the data previously reported at 60 GHz. These results show that the literature data vary significantly from one study to another depending on the sample type. Besides, since the skin consists of approximately 65% of free water [7], its complex permittivity is strongly dispersive and temperature-dependent; this should be also taken into account for definition of an accurate skin permittivity model.
To validate our measurement technique and obtain reference data for the skin-equivalent phantom, we performed a measurement campaign on a group of volunteers using two different techniques: open-ended coaxial slim probe [12] and a new method based on heating kinetics [13]. A very good agreement is demonstrated between our measurements and Gabriel [10] and Alekseev [11] data as shown in Fig. 3.
Comparison of our experimental result for the wrist skin permittivity (──) with Gabriel et al. (dry skin) (■) and Alekseev et al. (○) models. Error bars represent ±5% deviations around Gabriel’s reference values.
2.2. Numerical skin-equivalent phantom
Taking into account the very shallow penetration of millimeter waves into the skin (typically 0.5 mm at 60 GHz), using homogeneous skin-equivalent phantoms provides accurate results for the antenna / human body interaction evaluation as well as for the propagation channel characterization [28]. For the broadband analysis, dispersive models can be used. Debye model with a single relaxation time τ equal to that of free water at the same temperature was demonstrated to provide a good accuracy for modeling the experimental permittivity data in the considered frequency range [12]:
ε*=ε∞+Δε1+jωτ+σjωε0.E1
\n\t\t\t\t
In this equation, ω=2πf, f [Hz] is the frequency, Δε=εs-ε∞ is the magnitude of the dispersion of the free water fraction of skin, εs is the permittivity at ωτ<<1, ε∞ is the optical permittivity, ε0=8.85 10-12 F/m, and σ [S/m] is the ionic conductivity. The optimized parameters that fit to the measured permittivity in the 55-65 GHz range are the following: ε∞=4.1, εs=34.8, τ=6.9 × 10-12 s, and σ=0.7 S/m [12]. This model allows an accurate representation of typical broadband dielectric properties of dry skin in the numerical modeling.
2.3. Experimental skin-equivalent phantom
2.3.1. Composition
The main components employed for the fabrication of a homogeneous semi-solid skin-equivalent phantom are the following:
Deionized water. Water is the main constituent of the phantom because it is also the main skin component. It primarily determines the dispersive behavior of the phantom.
Agar. It is employed for the retention of self-shaping, and its contribution to the phantom dielectric properties is negligible for small concentrations (typically below 4%).
Polyethylene powder. It is used to tune the real and imaginary parts of the phantom permittivity.
TX-151. Since the agar and polyethylene powder cannot be mixed directly, the viscosity is increased using TX-151.
Sodium azide (NaN3). It serves as a preservative.
2.3.2. Fabrication procedure
The fabrication steps are the following. Deionized water, sodium azide, and agar are mixed in a kettle and heated on a stove, while the mixture is continuously stirred. When this liquid starts boiling, heating is stopped. TX-151 is sprinkled into the liquid and quickly mixed. Then the polyethylene powder is added into the stirred liquid. Finally, the obtained mixture is poured into a mold and cooled in the same container for a few hours to room temperature for solidification. Using alginate gel powder, molds with realistic body-specific shapes can be manufactured for the phantom fabrication as illustrated in Fig. 4.
Particular attention should be paid to the following critical points. First, to avoid variations of dielectric properties from one phantom to another, the room temperature should remain identical (in our case 20±1°C) during the fabrication and further measurements. Second, the type of polyethylene powder is important; we recommend using particles with an average diameter of 20µm and low density ~ 900–1100 kg/m3. Finally, to preserve the dielectric properties of the phantom over time, it is important to avoid water evaporation since this would result in a decrease of the permittivity. This can be for instance achieved by wrapping the phantom in a plastic film. More details regarding the phantom preparation procedure can be found in [12].
Figure 4.
Skin-equivalent phantom representing an arm and a hand: (a) fabrication of an alginate mold; (b) alginate mold of a human arm; (c) fabrication of the phantom liquid; (d) the phantom is extracted after being cooled inside the mold; (e) final result of a realistic human arm phantom.
2.3.3. Dielectric properties
The measured dielectric properties of the skin-equivalent phantom and skin are compared in Fig. 5. The dielectric properties of the proposed phantom are within ±10% of the measured skin permittivity. Table 2 compares the dielectric properties of the phantom measured using the coaxial probe and the heating kinetics technique [13] to those of the reference values provided by Gabriel et al. [10]. The measured data are in excellent agreement with the reference data and demonstrate that this phantom can be used for antenna measurement, on-body propagation, and dosimetric studies.
Figure 5.
Dielectric properties of the skin-equivalent phantom compared to those of skin [12]. Error bars represent ±10% of the measured skin permittivity.
\n\t\t
\n\t\t
\n\t\t
\n\t\t
\n\t\t
\n\t\t
\n\t\t\t
\n\t\t\t
\n\t\t\t\t\n\t\t\t\t\tε*\n\t\t\t\t\n\t\t\t
\n\t\t\t
\n\t\t\t\t\n\t\t\t\t\tΔε*\n\t\t\t\t\n\t\t\t
\n\t\t\t
\n\t\t\t\t\n\t\t\t\t\tR\n\t\t\t\t\n\t\t\t
\n\t\t\t
\n\t\t\t\t\n\t\t\t\t\tδ (mm)\n\t\t\t\t\n\t\t\t
\n\t\t
\n\t\t
\n\t\t\t
Reference value (Gabriel et al.) [10]
\n\t\t\t
7.98 – j10.9
\n\t\t\t
_
\n\t\t\t
0.38
\n\t\t\t
0.48
\n\t\t
\n\t\t
\n\t\t\t
Phantom (coaxial probe)
\n\t\t\t
7.4 – j11.4
\n\t\t\t
7.3% - j 4.6%
\n\t\t\t
0.39
\n\t\t\t
0.45
\n\t\t
\n\t\t
\n\t\t\t
Phantom (heating kinetics)
\n\t\t\t
8.3 – j10.8
\n\t\t\t
4% - j 0.9%
\n\t\t\t
0.38
\n\t\t\t
0.49
\n\t\t
\n\t
Table 2.
The dielectric properties of the proposed phantom (using two different techniques) compared to those of the reference data provided by Gabriel et al. [10]. Δε* is the error relative to Gabriel et al. data.
2.3.4. Validation
To further confirm the reliability of this phantom, we performed SAR measurement using a high-performance thermal imaging camera (FLIR SC500, FLIR Systems, Wilsonville, OR, USA) and the measurement set-up shown in Fig. 6a. The SAR assessment methodology is described in Fig. 6b. The temperature dynamic, recorded using the IR camera, is fitted to the one-dimensional bio-heat transfer equation [12]. The fitting procedure is performed by minimizing the standard deviation value varying the incident power density (IPD). Once the IPD value has been determined, the SAR can be found (Fig. 6b). The simulated and measured SAR results are in excellent agreement (Fig. 6) which confirms the accuracy of this phantom.
Figure 6.
(a) SAR measurement set-up. (b) SAR assessment methodology from the temperature rise. (c) Simulated and measured SAR results.
3. Antennas for off-body communications at millimeter-waves
At microwaves, it is widely accepted that antennas placed in close proximity to a lossy medium experience strong power absorption, radiation pattern distortion, shift in resonance frequency, and changes in the input impedance, e.g. [1],[19]-[21]. Therefore, when placed close to the human body, wearable antennas need to be designed to operate in a robust way so that the influence of the body on the antenna performance is minimized. Patch antennas have been identified as one of the best solutions for off-body communications [1]. These are simple and low-cost structures, and their radiation at broadside allows maximizing radiations at the opposite side of the human body while reducing radiation towards the body.
At millimeter waves, the electromagnetic coupling between antennas and the human body as well as possible perturbations of antenna characteristics due to the body remain almost unexplored. In addition, in this frequency range, a particular attention must be paid to the power absorbed in the body since this absorption is very localized.
In this Section, the interactions between the human body and millimeter wave antennas, optimized for off-body communications, are studied numerically and experimentally. First, requirements for wearable antennas for off-body communications are briefly outlined. Then, the influence of the antenna feeding is investigated. Then, a four-patch antenna array is designed and characterized numerically and experimentally both in free space and on the skin-equivalent phantom described in the previous section. SAR and incident power density distributions on the phantom are determined using the methodology presented in [12]. Finally, in order to study the capabilities of the integration into textiles, a similar four-patch antenna array is designed and fabricated on a fabric.
3.1. Antenna requirements for off-body communications
Wearable antennas have to be as compact as possible to be integrated with the transceiver. They have to be efficient with minimal power absorption inside the human body that behaves as a highly lossy dispersive dielectric material at millimeter waves. The antennas also have to be light weight and, in some particular cases, conformable to the human body shape. Because of the high atmospheric attenuation at 60 GHz and limitations on the radiated power, medium-gain antennas (~12dBi) are often required [14]. Indeed, in controlled environments, line-of-sight (LOS) channels can be efficiently exploited using medium-gain passive antennas, whereas directive beam steering antennas are desirable for non-line-of-sight (NLOS) channels so as to comply with the power link budgets [14]-[18]. In our studies, we only consider LOS scenarios and thus restrict our consideration to passive medium-gain antennas.
3.2. Influence of the antenna feeding
The influence of the antenna feeding is investigated when the antenna is placed on the human body. At lower frequencies, patch antennas have been presented as the best solution for off-body communications. However, at millimeter waves, the influence of spurious waves due to the feeding lines on radiating patterns cannot be neglected. That is why, multilayer antenna designs are generally considered in order to overcome this issue.
The interaction with the human body and two types of patch antennas is studied: (1) a linearly-polarized antenna and (2) a linearly-polarized aperture coupled patch antenna. These antennas are printed on a 0.127mm-thick RT Duroid 5880 substrate (h=127 μm, εr=2.2, tanδ=0.003).
3.2.1. Microstrip patch antenna
A simple patch antenna is optimized to achieve a maximum gain at 60 GHz. The dimensions are given in Fig. 7. The reflection coefficient and radiation patterns are studied numerically in free space and on the human body when the antenna is placed at 1mm above the phantom. For the numerical modeling, a parallelepipedic 10 × 100 × 100 mm3 phantom is used and a Debye model has been used to express the complex permittivity ε* of the skin-equivalent phantom (see Section 2).
The reflection coefficient is very slightly affected by the human body (Fig. 8) and the radiation pattern remains stable at the opposite side of the human body, whereas the backward radiations are highly reduced in H-plane (Fig. 9). These results demonstrate that microstrip patch antennas are only slightly sensitive to the human body proximity at 60 GHz.
Figure 7.
Microstrip patch antenna at 60 GHz.
Figure 8.
Simulated reflection coefficient of the microstrip patch antenna. ── In free space. ─ ─ On the skin-equivalent phantom.
Figure 9.
Simulated radiation pattern of the microstrip patch antenna. ── In free space. ─ ─ On the skin-equivalent phantom.
3.2.2. Aperture coupled patch antenna
Fig. 10 shows the configuration of the aperture-coupled patch antenna (ACPA). The slot is optimized to 0.26 × 1 mm² for maximum coupling with a stub length of 0.34 mm. In order to consider the easiness of implementation, a 0.2-mm-thick ground plane is employed. The antenna consists of a patch with optimized dimension of 1.33×1.24 mm² on a 0.127-mm-thick RT Duroid 5880 substrate. Low thickness and low-permittivity substrate are used for reducing surface waves.
The reflection coefficient S11 (Fig. 11) and radiation patterns (Fig. 12) are investigated in free space and on the skin-equivalent phantom (antenna/body spacing h=1mm). It is worthwhile to note that the S11 is even less affected by the human body proximity for the ACPA. However, as far as the radiation pattern is concerned, the backward radiations are highly reduced (i.e. by at least 10 dB). This demonstrates that absorptions inside the body are higher for the ACPA and the SAR should be carefully studied. The gain is very slightly increased on the human body (Table 3).
Figure 10.
Aperture coupled patch antenna. (a) 3D and (b) 2D schematic representation of the antenna model and dimensions.
Figure 11.
Simulated reflection coefficient of the aperture coupled patch antenna. ── In free space. ─ ─ On the skin-equivalent phantom.
Figure 12.
Simulated radiation pattern of the aperture coupled patch antenna. ── In free space. ─ ─ On the skin-equivalent phantom.
3.2.3. Specific Absorption Rate (SAR) comparison
The SAR are compared for the microstrip patch antenna and ACPA for an antenna/body spacing h=1mm and for an incident power of 1W. The peak SAR obtained for the ACPA is 41 times higher compared to that obtained with the microstrip patch antenna. Therefore, it is not recommended to use ACPA because the input power would be highly limited compared to that of the microstrip patch antenna to comply with the exposure limits [22], resulting in a lower link budget (gain and efficiency remain almost equivalent for both antennas).
\n\t\t
\n\t\t
\n\t\t
\n\t\t
\n\t\t
\n\t\t
\n\t\t\t
\n\t\t\t
\n\t\t\t\tMicrostrip patch antenna\n\t\t\t
\n\t\t\t
\n\t\t\t\tACPA\n\t\t\t
\n\t\t
\n\t\t
\n\t\t\t
\n\t\t\t
\n\t\t\t\tFree space\n\t\t\t
\n\t\t\t
\n\t\t\t\tOn the phantom\n\t\t\t
\n\t\t\t
\n\t\t\t\tFree space\n\t\t\t
\n\t\t\t
\n\t\t\t\tOn the phantom\n\t\t\t
\n\t\t
\n\t\t
\n\t\t\t
Peak SAR (W/kg)1\n\t\t\t
\n\t\t\t
-
\n\t\t\t
279
\n\t\t\t
-
\n\t\t\t
11485
\n\t\t
\n\t\t
\n\t\t\t
Peak gain (dBi)
\n\t\t\t
6.01
\n\t\t\t
6.03
\n\t\t\t
6.22
\n\t\t\t
6.70
\n\t\t
\n\t\t
\n\t\t\t
Efficiency (%)
\n\t\t\t
79.31
\n\t\t\t
74.44
\n\t\t\t
84.58
\n\t\t\t
77.67
\n\t\t
\n\t
Table 3.
Peak SAR, gain, and efficiency for the microstrip patch antenna and ACPA. 1For an incident power of 1W
3.2.4. Conclusion
Two patch antennas have been compared numerically in free space and on a skin-equivalent phantom. For the microstrip antenna and ACPA, the influence of the human body is very weak, and their performances remain stable. However, the SAR resulting from the ACPA is 41 times higher compared to that obtained with the microstrip antenna. Therefore, it is highly recommended to avoid aperture coupled feeds. If it is necessary, the feeding line could be sandwiched between two substrates with top and bottom grounds [23].
3.3. Patch antenna array
3.3.1. Antenna model
To satisfy the criteria summarized in Section 3.1 and following the conclusions drawn in Section 3.2, a microstrip-fed four-patch single-layer antenna array has been chosen [24]. It is printed on a thin RT Duroid 5880 substrate (h=127 μm, εr=2.2, tanδ=0.003). The layout is represented in Fig. 13a. A single rectangular patch antenna typically provides a 7 dBi gain; a 2×2 antenna array is chosen here to reach a gain of 12 dBi with about the same beamwidth in E-and H-planes. The inter-element spacing is selected to achieve a good trade-off between high gain and low side lobes. Similar 2×2 antenna arrays have already been reported in a multilayer configuration [25] or fed by a coaxial probe [26],[27], which would make them difficult to fabricate on flexible or textile substrates. Hence, here all patches are fed using a single-layer corporate feed network. The antenna is linearly-polarized along y-direction, and, for measurement purposes, it is mounted on a 3 mm-thick ground plane (Fig. 13b) to avoid significant substrate bending and to achieve an accurate and stable placement of a V-connector. In the future BAN applications, this kind of antennas is expected to be directly integrated into the clothing or wearable devices.
Figure 13.
Patch single-layer antenna array at 60 GHz [24]. (a) Schematic representation of the antenna model and dimensions. (b) Manufactured antenna with a V-connector.
3.3.2. Antenna performance
The antenna reflection coefficient S11 was measured in free space and on the skin-equivalent phantom (Fig. 14). It remains below-10 dB from 59 GHz to 65 GHz. It is clear that the skin-equivalent phantom does not affect the antenna reflection coefficient. This is not the case at microwaves where a resonance shift is usually observed.
In addition, the radiation patterns in E-and H-planes are plotted in Fig. 15 at 60 GHz. The gain was measured by the comparison method with a 20-dBi standard horn, and the directivity is found from a 3D radiation pattern measurement. It can be seen in Fig. 15 that front radiations remain very slightly affected.
The backward radiation was measured separately in both configurations. Whereas the measured level on the phantom is mainly reduced in the E-plane (Fig. 15c) due to the absorption and reflection, it remains very slightly affected in the H-plane (Fig. 15d). This could be expected since the absorption is higher when the E-field is parallel to the phantom surface [28]. These observations are in agreement with the calculated and measured SAR and incident power density (IPD) as shown in Fig. 16. More details regarding the measurement methodology can be found in [12].
At this frequency, the measured gains in free space and on the phantom equal 11.8 dBi (±0.3dB) and 11.9 dBi (±0.3dB), respectively. This demonstrates the small effect of the phantom presence. At 60 GHz, the measured directivity was assessed to be equal to 13.9 dBi (±0.3dB) and 14.1 dBi (±0.3dB), respectively. Comparison of the measured directivities with the measured gains leads to antenna efficiencies of 62% and 60%, respectively. This efficiency value is typical in V-band for this kind of antennas and could be further improved, for instance using a fused quartz substrate [25] instead of RT Duroid 5880. Whereas the antenna efficiency at microwaves can be strongly affected by the body presence even for patch antennas [1], it is found here that it remains stable in V-band.
Figure 14.
Measured reflection coefficient of the antenna. ——– In free space. ××× On the skin-equivalent phantom.
Figure 15.
Measured normalized radiation patterns at 60 GHz in E-and H-planes. ── Measurement in free space. ---Measurement on the phantom.
Figure 16.
SAR and IPD distributions at 60 GHz. (Left) Numerical results for the antenna on the skin. (Right) Measurements on the skin-equivalent phantom. Pin=322 mW.
3.4. Textile antennas
Textile antennas at millimeter waves could be of great interest for many applications. However, on-textile fabrication process is very challenging at these frequencies, especially due to the roughness of the textile surface and the size of textile fibers and electrotextiles with respect to the geometrical dimensions of the metallic patterns.
It was demonstrated in [29] that commercial textiles can be used as antenna substrates at millimeter waves. Some results are presented here showing a 60-GHz textile-based antenna for off-body wireless communications with the ability to be bent and deformed into an arbitrary shape. A simple, but representative patch antenna array is fabricated using an ad-hoc manufacturing process. Compared to the antenna presented in Section 3.3, this results in a highly flexible antenna.
3.4.1. Technological fabrication process
The fabrication process of millimeter-wave textile antennas has been presented in [29] and [31] (Fig. 17). The first step (Fig. 17a) consists in placing an electrotextile layer (e.g. ShieldIt Super) on the lower side of the textile (ground plane), and flexible copper foil on the top side. The second step (Fig. 17b) consists in micromachining the copper foil using a laser machine with optimized laser parameters to avoid any damage of the textile substrate such as ragged or burnt edges.
Hence, using a laser machine (ProtoLaser S, LPKF, OR, USA) operating at 1064nm with a pulse duration of 7.5ns and a spot size equals 25 μm, the laser parameters were optimized. A laser fluence of 24.4 mJ/cm2 with three cycles on the surface of the substrate has been used for the copper foil ablation (repetition rate=75 kHz, power=16.0 W) without affecting the textile substrate. These fabrication conditions lead to a geometrical accuracy of about 10 μm. It is worthwhile to underline that the accuracy reported so far with two conductive fabrics, namely knitted P130 and woven Nora fabric, is only about ±0.5mm and ±0.15mm, respectively [32]. Finally, the last step (Fig. 17c) consists in manually removing the unwanted parts of the copper foil from the surface of the textile.
Figure 17.
Main technological steps for the manufacturing of printed circuits and antennas on textiles in V-band.
Whereas in most fabrication processes reported so far, the metallic part is cut separately and then adhered to the dielectric layer, cutting out the desired pattern directly on the dielectric layer avoids additional discrepancies. Example of manufactured microstrip antennas and lines are shown in Fig. 18. The devices are very flexible and the pattern quality (dimensions, sharpness of the edges) is very satisfactory (Fig. 18).
Figure 18.
Examples of fabricated textile antennas and microstrip lines.
3.4.2. Textile characterization
The choice of a substrate thickness, dielectric constant εr and loss tangent tanδ is essential when it comes to millimeter waves in order to avoid losses and also to enhance the efficiency. The methodology employed to retrieve the dielectric properties of any textile layer is explained here. As an example, this methodology is applied to a 0.2 mm-thick cotton woven fabric extracted from a shirt. Its permittivity and loss tangent are determined in V-band as explained below. The devices under test have been manufactured using the fabrication process described in Section 3.4.1.
The characterization technique is simple and straightforward and consists in two parts:
First, the relative permittivity is retrieved using the open-stub technique. To this end, we have designed a transmission line loaded by an open-ended parallel stub (Fig. 18b) whose length ls is chosen to provide a resonance close to 60 GHz. The resonant frequency is measured in transmission with a V-band Anritsu universal test fixture 3680 V (Fig. 18b) connected to an Agilent 8510XF vector network analyzer (VNA). The measurement set-up has been calibrated using a full 2-port calibration procedure. The measured transmission coefficient S21 is represented in Fig. 19a (solid line) from 10 to 65 GHz. In simulations, the relative permittivity of the textile is tuned numerically until the theoretical S21 curve coincides with the measured one.
Second, the loss tangent is estimated through a differential measurement in transmission of two matched 50-Ω microstrip lines of different lengths (Fig. 18c). This enables determination of the total insertion loss (Fig. 19b), and tanδ is found by fitting the measured and simulated data. Our experimental data show that the insertion loss of transmission lines fabricated on cotton woven fabric reaches about 1.6 dB/cm at 60 GHz, which is larger than values obtained with conventional substrates [29].
The best agreement between simulations and experiments is obtained with εr=2.0 and tanδ=0.02. These values will be used for the antenna design. Since commercial textiles are lossy, a slight deviation in the determination of their loss-tangent would have a minor impact. Therefore, deviations due to the use of electromagnetic software are considered as acceptable.
The insertion loss of a 50-Ω microstrip line printed on a 0.2mm-thick textile is about 1.6 dB/cm, which is quite important compared to conventional substrates such as RT Duroid 5880, fused quartz and alumina [29]. However, these substrates are not as flexible as textiles. For a fair comparison, we should consider a flexible substrate such as PDMS where the insertion losses are much more important (~3 dB/cm for a 0.2mm-thick PDMS) [30].
Figure 19.
a) Transmission coefficient S21 of the stub loaded microstrip line (ls=4.58 mm, L=50 mm. (b) Insertion loss of a 50 Ω line. The numerical data assume εr=2.0 and tanδ=0.02. Measured (——) and computed (– –) data.
3.4.3. Microstrip patch antenna
The fabricated textile patch antenna operating at 60 GHz is shown in Fig. 20a. For measurement purpose, it is integrated with a V-connector. The flexibility of the antenna is demonstrated in Fig. 20b. The antenna was optimized to operate at 60 GHz using CST Microwave Studio. The reflection coefficient and radiation pattern of the textile antenna have been characterized in free space and on a parallepipedic skin-equivalent phantom (10×100×100 mm3). The complex permittivity of the phantom equals that of human skin within the maximum error of 10% in the 57-64 GHz range [12].
First, the simulated and measured reflection coefficient is represented in Fig. 21. A frequency shift of only 2.5% is observed between computed and simulated results. It could be due to a change in the substrate permittivity and under-or over-etching of the microstrip line. Whereas at microwave frequencies patch antennas experience shift in resonance frequency [1], it can be seen that the reflection coefficient of the proposed antenna is immune from the human body proximity.
Measured radiation patterns in free space and on the homogeneous phantom were measured at 60 GHz. It was observed that the radiation pattern is very slightly affected by the phantom. The simulated and measured gains equal 4.3 dBi and 4.2 dBi, respectively. On the phantom, the maximum gain is decreased by 0.2 dB and 0.7dB in simulation and in measurement, respectively. Hence, whereas at microwaves patch antennas could be highly affected in terms of gain and efficiency [1], at millimeter waves the antenna performances remains unchanged.
Figure 20.
Photography of the fabricated patch antenna with a V-connector.
Figure 21.
Reflection coefficient of the wearable patch antenna. ── Computed result in free space.── Measured result in free space. —■—Measured result on the skin-equivalent phantom.
3.4.4. Microstrip patch antenna array
A microstrip-fed four-patch single-layer antenna array printed on the 0.2mm-thick textile has been designed (Fig. 22a) [29]. The array is fed by a 15.2mm-long microstrip line to avoid too strong reflections from the V-connector (Fig. 22). In practice, as textiles are more lossy than classical substrates, it is recommended to reduce the access line length as much as possible. Whereas the antenna could be fed using a central probe, (as shown in Fig. 22b), the microstrip feed line is the easiest solution to perform measurements on textile. We will discuss the impact of this microstrip line in terms of loss and distortion of the radiation pattern. The fabricated antenna integrated with a V-connector is shown in Fig. 23.
Its reflection coefficient S11 is measured using a 110-GHz Agilent 8510XF VNA and is shown in Fig. 24. Excellent agreement is obtained between simulated and measured results. The reproducibility of these results has been demonstrated and more information can be found in [29].
Figure 22.
Layout of the microstrip antenna array printed on textile. (a) Antenna fed using a long microstrip line. (b) Antenna fed using a central coaxial probe.
Figure 23.
Measurement set-up on the skin-equivalent phantom for a distance between the ground plane and the phantom equal to d=0 mm.
Figure 24.
Reflection coefficient of the microstrip antenna array printed on textile. —□— Measured.---Simulated.
The radiation patterns in E-and H-planes were measured in IETR’s millimeter-wave anechoic chamber. The gain was measured by the comparison method with a 20-dBi standard horn, and the directivity is found from a 3D radiation pattern measurement. The co-polarization components measured in E-and H-planes at 60 GHz are in a good agreement with the computed ones (Fig. 25). In E-plane, the non-symmetry of the co-polarization component is attributed to the spurious radiation of feeding lines whose width is larger compared to standard substrates at millimeter waves like RT Duroid 5880 (see Section 3.3), or other commonly used substrate such as fused quartz or Alumina. The main characteristics of these three different substrates are compared in [29] with those of the textile used here. These data show that textile exhibits higher loss and that feeding lines are larger.
The simulated cross-polarization level remains lower than –20 dB at broadside in E-and H-planes. As expected and as already noticed in many previous papers (e.g. [24]), the measured values are much higher due to reflections and scattering on the V-connector and metallic support (Fig. 25b).
Besides, simulations have shown that the V-connector also affects the gain and directivity; therefore, for comparison purpose, these results are given for both configurations (i.e. with and without connector). The cross-polarization level could be further improved using a multilayer antenna design, e.g. [25]. However, as explained in Section 3.2, the latter is not recommended for on-body applications due to the relatively high SAR levels.
The effect of the central microstrip line exciting the antenna array has been investigated numerically comparing the radiation patterns of the proposed array (Fig. 22a) and those of a coaxial-fed array (Fig. 22b) [29]. These results (not shown here) demonstrate that the increase of the cross-polarization levels and side lobe levels in E-plane is due to the main feed line.
In addition, the gain, directivity and efficiency of these two antennas have been characterized (Table 4). High losses are experienced in the feed line (about 3.3 dB). In order to increase the antenna gain and efficiency, the feed line could be shortened or even suppressed (Fig. 22b).
Finally, the antenna performance (i.e. reflection coefficient and radiation) was tested after a number of hand washing cycles. The antenna was measured before and after washing when fully dried; its performance remained unchanged. However, to extend the life duration of the antenna, the authors would recommend waterproofing the whole antenna.
\n\t\t
\n\t\t
\n\t\t
\n\t\t
\n\t\t
\n\t\t
\n\t\t
\n\t\t
\n\t\t\t
\n\t\t\t
\n\t\t\t\tGain (dBi)\n\t\t\t
\n\t\t\t
\n\t\t\t\tDirectivity (dBi)\n\t\t\t
\n\t\t\t
\n\t\t\t\tEfficiency (%)\n\t\t\t
\n\t\t
\n\t\t
\n\t\t\t
\n\t\t\t
\n\t\t\t\tSim.\n\t\t\t
\n\t\t\t
\n\t\t\t\tMeas.\n\t\t\t
\n\t\t\t
\n\t\t\t\tSim.\n\t\t\t
\n\t\t\t
\n\t\t\t\tMeas.\n\t\t\t
\n\t\t\t
\n\t\t\t\tSim.\n\t\t\t
\n\t\t\t
\n\t\t\t\tMeas.\n\t\t\t
\n\t\t
\n\t\t
\n\t\t\t
Microstrip-fed array (Fig. 22a)
\n\t\t\t
8.6
\n\t\t\t
8.0
\n\t\t\t
12.1
\n\t\t\t
11.9
\n\t\t\t
45
\n\t\t\t
41
\n\t\t
\n\t\t
\n\t\t\t
Coxial-fed array (Fig. 22b)
\n\t\t\t
11.9
\n\t\t\t
-
\n\t\t\t
13.1
\n\t\t\t
-
\n\t\t\t
75
\n\t\t\t
-
\n\t\t
\n\t
Table 4.
Comparison of antenna performances in terms of gain, directivity and efficiency.
Figure 25.
Normalized radiation patterns in co-and cross-polarization at 60 GHz. ── Simulation in free space. —■— Measurement in free space. --- Measurement on a skin-equivalent phantom.
3.5. Conclusion
Based on computed and measured results, antennas operating at millimeter-waves are very slightly sensitive to the human body. Besides, guidelines regarding the type of antennas, minimizing the interactions with the body, are provided. The feeding of the antenna is a critical point and aperture-coupled microstrip line-fed patch antennas should be avoided since it results in significantly higher body absorptions. A good alternative would be to use an aperture-coupled stripline-fed patch antenna instead.
Finally, textile antennas at millimeter-wave have been demonstrated with encouraging results. The textile can be accurately characterize and employed as antenna substrate. The textile antenna prototypes, fabricated using a simple and commercially compatible fabrication process, demonstrate excellent flexibility capabilities which would simplify the integration in clothes.
4. Antennas for on-body communications at millimeter waves
Whereas off-body communications appear to be a good solution at millimeter waves, on-body communications might be more challenging. In particular, significant shadowing effect from the human body is expected to make non-line-of-sight communications very difficult if not impossible. In [34], an on-body scenario has been numerically investigated in terms of propagation and demonstrates that short-range communications are achievable. The propagation issues are out of the scope of this Chapter; the readers can refer to the following papers for more details [33]-[35]. A few antennas optimized for on-body communications have been presented in the literature so far [31],[36],[37]. This Section will emphasize on the antenna performances in close proximity to the body.
4.1. Antenna requirements for on-body communications
On-body antennas should be as compact as possible to be integrated with a transceiver. As for off-body antennas, they have to be light weight and possibly conformable to the human body shape. Because of the high attenuation related to the propagation on a lossy dielectric (i.e. human body), medium-gain antennas (~12dBi) are required. The radiation pattern should be maximized toward the direction of propagation to minimize losses and make end-fire antennas excellent solutions. As the power is directed toward the body surface, absorptions inside the human is of uppermost concern.
4.2. End-fire antenna
A compact planar and flexible Yagi-Uda antenna covering the 57-64 GHz range designed for on-body communications is presented. The antenna is characterized in free space in terms of reflection coefficient, radiation pattern, and efficiency. The effect of the human body on the antenna characteristics is studied numerically and experimentally using a skin-equivalent phantom. The antenna performances are also studied under bending conditions. An on-body scenario is numerically investigated in terms of propagation.
4.2.1. Antenna model
High gain antenna is required for a line-of-sight path of human body dimensions. Furthermore, the maximum of the radiation pattern should be tangential to the body surface in order to reduce radiation off the body and thus minimizing interference among different BANs. Hence, a low-profile high-gain antenna with an end-fire radiation pattern printed on a 0.254mm-thick RT Duroid 5880 substrate (εr=2.2, tanδ=0.003) is proposed. The layout is represented in Fig. 26. For measurement purpose the antenna prototype is mounted with a V-connector (Fig. 27).
Figure 26.
Layout of the printed Yagi-Uda antenna. Dimensions are in mm. (a) Three dimensions view. (b) Top layer.
Figure 27.
Manufactured antenna with a V-connector.
4.2.2. Antenna performance in free space
The reflection coefficient S11 of the antenna array is measured with a 110 GHz vector network analyzer (Agilent 8510XF) using a V-connector (Fig. 27). The measured and simulated S11 (Fig. 28) are below-10 dB in the whole 57-64 GHz range. The numerical model does not include the V-connector.
The radiation patterns in E-and H-planes are plotted in Fig. 29. The simulated and measured radiation patterns at 60 GHz are in good agreement. The cross-polarization remains lower than-14 dB in the E-and H-planes at broadside. The gain was measured by the comparison method with a 20-dBi standard horn. At this frequency, the measured and computed gains equal 11.8 dBi and 12.1 dBi respectively. The losses of the V-connector (~0.8 dB) at 60 GHz were measured separately and taken out.
The antenna efficiency defined as the measured gain over the computed directivity equals to 86% at 60 GHz. It is in agreement with the simulated efficiency which equals 92%.
Figure 28.
Measured and simulated reflection coefficient of the antenna in free space. --- Measurement. ── Simulation.
Figure 29.
Measured and simulated radiation patterns in free space at 60 GHz in (a) E-and (b) H-planes. —■— Measured co-pol. ── Computed co-pol. ── Measured cross-pol.
4.2.3. Antenna under bending conditions
As it is difficult to keep the antenna flat in wearable applications, antenna performances under bending conditions is an important factor to be examined. The reflection coefficient and the H-plane radiation pattern are investigated when the antenna is placed on semi-cylindrical Rohacell HF51 foam with a radius R of 15mm (Fig. 30).
The chosen radius represents extremely severe test. The S11 is measured when the antenna is bent in the H-plane. The S11 remains below-10 dB in the whole 57-64 GHz range (Fig. 31).
Measured and simulated radiation pattern for H-plane bending are represented in Fig. 32. The maximum radiation follows the direction of the directors (-46°). Besides, the measured gain equals 11.1 dBi. This is in good agreement with the simulated gain (11.0 dBi). Compared to the gain in free space, a drop of 0.7 dB is observed in measurement.
Figure 30.
Bending antenna in the H-plane placed on a semi-cylindrical foam with R=15mm.
Figure 31.
Measured reflection coefficient of the bent antenna mounted on semi-cylindrical foam. --- Flat. ── R=15mm.
Figure 32.
H-plane radiation pattern of the bent antenna (R=15mm) mounted on a semi-cylindrical foam. —■— Measured co-pol. ---- Computed co-pol.
4.2.4. Antenna performances on the human body
The antenna characteristics are assessed when placed on a skin-equivalent phantom (Fig. 33) in terms of reflection coefficient, radiation pattern, gain, and efficiency. The measured reflection coefficients of the antenna mounted on the phantom at different antenna/body spacing h are compared to that obtained in free space in Fig. 34. For h=5mm, the reflection coefficient is very slightly affected. For h=2mm, even though the S11 is much more affected and a frequency shift is observed, it remains below-10dB within the whole 57-64 GHz.
Figure 33.
Antenna on the skin-equivalent phantom.
The measured radiation patterns in both E-and H-planes at 60 GHz of the antenna placed on the skin-equivalent phantom are represented in Fig. 35 for h=5.6mm and h=2mm. Both E-and H-planes are strongly affected by the human body because of reflection on and absorption in the body.
Here, the radiation pattern is titled because of reflections occurring at the air/phantom interface. A tilt of 10° and 21° is observed for an antenna/body spacing of 5.6mm and 2mm, respectively. The simulated and measured gains and the simulated efficiency are summarized in Table 5 for different antenna/body spacing. The efficiency decreases with h. However, the maximum gain of the antenna increases on the phantom (up to 3dB increase for h=5.6mm). Compared to the free space configuration, radiations toward the human body are significantly reduced because of reflections from and absorptions in the human body. Hence, when the antenna is mounted on the phantom, its performance remains satisfactory in terms of reflection coefficient, radiation pattern, and efficiency.
Figure 34.
Measured reflection coefficient of the antenna array on the homogeneous phantom. — Free space. ▪▪▪▪▪ On phantom with h=5.6mm.---On phantom with h=2mm. — On phantom with h=0mm.
Figure 35.
Measured and simulated radiation patterns on the skin-equivalent phantom at 60 GHz. —■— Measured co-pol. ── Computed co-pol. ── Measured cross-pol.
\n\t\t
\n\t\t
\n\t\t
\n\t\t
\n\t\t
\n\t\t\t
\n\t\t\t\tAntenna/phantom separation h (mm)\n\t\t\t
\n\t\t\t
\n\t\t\t\tGain (dBi)\n\t\t\t
\n\t\t\t
\n\t\t\t\tSimulated efficiency (%)\n\t\t\t
\n\t\t
\n\t\t
\n\t\t\t
\n\t\t\t\tSimulated\n\t\t\t
\n\t\t\t
\n\t\t\t\tMeasured\n\t\t\t
\n\t\t
\n\t\t
\n\t\t\t
\n\t\t\t\t∞\n\t\t\t
\n\t\t\t
12.1
\n\t\t\t
11.8
\n\t\t\t
92.2
\n\t\t
\n\t\t
\n\t\t\t
5.6
\n\t\t\t
15.1
\n\t\t\t
15.2
\n\t\t\t
74.9
\n\t\t
\n\t\t
\n\t\t\t
2
\n\t\t\t
13.6
\n\t\t\t
13.6
\n\t\t\t
68.2
\n\t\t
\n\t
Table 5.
Antenna gain and efficiency for different antenna/body spacing.
4.3. Conclusions
A compact planar Yagi-Uda antenna covering the 57-64 GHz range has been designed for on-body communications. The effect of the human body on the antenna characteristics has been studied numerically and experimentally using a skin-equivalent phantom. It was shown that the distance between the antenna and the human body has a strong impact on the antenna performances. The antenna was also studied under bending conditions demonstrating satisfactory performances. The same antenna has been successfully optimized and fabricated on textile [31].
5. Conclusion
Challenges and progress in antennas and their interaction with the human body in body-centric scenarios at millimeter-wave frequencies have been presented in this Chapter. Recent progress in manufacturing and modeling experimental phantoms has been discussed. These phantoms play a key role in characterizing the antenna performance in close proximity to the human body.
As far as off-body communications are concerned, it was shown that the feeding type is an important factor since it can strongly influence absorption in the human body. In addition, performances of patch antenna arrays in close proximity to the human body have been evaluated showing very slight impact on the antenna performance. Besides, a textile patch antenna array, operating at millimeter waves, was successfully demonstrated using a commercially-available textile. An accurate and low-cost fabrication process has been introduced. Research work should now be focused on the interconnections between textile antennas and Radio Frequency Integrated Circuits (RFIC) since this issue has not been tackled yet.
Finally, as end-fire antennas appear to be the best solution for on-body communications, a Yagi-Uda antenna has been investigated. It appears that the antenna radiation pattern is strongly affected by the separation between the antenna and the human body. This antenna is robust against bending which is an important asset if this antenna would be implemented on textile as shown in [31]. Other antenna designs for on-body communications were introduced in [38].
While these results are promising, millimeter-wave wireless systems still have considerable challenges to overcome to enable mass commercialization. First, mm-wave wireless must address challenging RF impairments such as fast fading and delay spread conditions making demodulation and equalization particularly difficult with reasonable architectures and complexities. Second, millimeter-wave transceivers require giga-samples per second (GS/s) scale data-converters with considerable resolutions leading to high power consumption (even in advanced technology nodes). Finally, mm-wave schemes must prove themselves competitive with advanced and adaptive modulation and channel coding schemes (256 QAM and beyond) like 802.11ac 5th generation WiFi that can also reach high data rates (6.77 Gbit/s nominal) while being built upon existing wireless hardware and infrastructure in the 5.83 GHz ISM band.
Acknowledgments
This work was supported by French National Research Agency (ANR) under Grant ANR-09-RPDOC-003-01 (Bio-CEM project), by Labex CominLabs (ANR program "Investing for the Future" ANR-10-LABX-07-01) and Brittany Region under ResCor/BoWi project and by National Center for Scientific Research (CNRS), France. Part of this work was performed using HPC resources from GENCI-IDRIS (grant 2013-050779).
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Electromagnetic properties and modeling of the human body",level:"1"},{id:"sec_2_2",title:"2.1. Interaction of millimeter waves with the human body",level:"2"},{id:"sec_3_2",title:"2.2. Numerical skin-equivalent phantom",level:"2"},{id:"sec_4_2",title:"2.3. Experimental skin-equivalent phantom",level:"2"},{id:"sec_4_3",title:"2.3.1. Composition",level:"3"},{id:"sec_5_3",title:"2.3.2. Fabrication procedure",level:"3"},{id:"sec_6_3",title:"Table 2.",level:"3"},{id:"sec_7_3",title:"2.3.4. Validation",level:"3"},{id:"sec_10",title:"3. Antennas for off-body communications at millimeter-waves",level:"1"},{id:"sec_10_2",title:"3.1. Antenna requirements for off-body communications",level:"2"},{id:"sec_11_2",title:"3.2. Influence of the antenna feeding",level:"2"},{id:"sec_11_3",title:"3.2.1. Microstrip patch antenna",level:"3"},{id:"sec_12_3",title:"3.2.2. Aperture coupled patch antenna",level:"3"},{id:"sec_13_3",title:"Table 3.",level:"3"},{id:"sec_14_3",title:"3.2.4. Conclusion",level:"3"},{id:"sec_16_2",title:"3.3. Patch antenna array",level:"2"},{id:"sec_16_3",title:"3.3.1. Antenna model",level:"3"},{id:"sec_17_3",title:"3.3.2. Antenna performance",level:"3"},{id:"sec_19_2",title:"3.4. Textile antennas",level:"2"},{id:"sec_19_3",title:"3.4.1. Technological fabrication process",level:"3"},{id:"sec_20_3",title:"3.4.2. Textile characterization",level:"3"},{id:"sec_21_3",title:"3.4.3. Microstrip patch antenna",level:"3"},{id:"sec_22_3",title:"Table 4.",level:"3"},{id:"sec_24_2",title:"3.5. Conclusion",level:"2"},{id:"sec_26",title:"4. Antennas for on-body communications at millimeter waves",level:"1"},{id:"sec_26_2",title:"4.1. Antenna requirements for on-body communications",level:"2"},{id:"sec_27_2",title:"4.2. End-fire antenna",level:"2"},{id:"sec_27_3",title:"4.2.1. Antenna model",level:"3"},{id:"sec_28_3",title:"4.2.2. Antenna performance in free space",level:"3"},{id:"sec_29_3",title:"4.2.3. Antenna under bending conditions",level:"3"},{id:"sec_30_3",title:"Table 5.",level:"3"},{id:"sec_32_2",title:"4.3. Conclusions",level:"2"},{id:"sec_34",title:"5. Conclusion",level:"1"},{id:"sec_35",title:"Acknowledgments",level:"1"}],chapterReferences:[{id:"B1",body:'P. S. Hall and Y. Hao, “Antennas and propagation for body centric communications systems,” Artech House, Norwood, MA, 2006, ISBN-10: 1-58053-493-7.'},{id:"B2",body:'D. Guha and Y. M. M. Antar, “Microstrip and printed antennas: new trends, techniques and applications,” Wiley-Blackwell, 2011, ISBN-10: 0470681926.'},{id:"B3",body:'T. Baykas, C. S. Sum, Z. Lan, J. Wang, M. A. Rahman, and H. Harada, “IEEE 802.15.3c: the first IEEE wireless standard for data rates over 1 Gb/s,” IEEE Communications Mag., vol. 49, no. 7, pp. 114–121, Jul. 2011.'},{id:"B4",body:'S. L. Cotton, W. G. Scanlon, and P. S. Hall, “A simulated study of co-channel inter-BAN interference at 2.45 GHz and 60 GHz,” Europ. Wirel. Techn. 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Zammit",slug:"joseph-a.-zammit"}]},{id:"14717",title:"Electrically Small Microstrip Antennas Targeting Miniaturized Satellites: the CubeSat Paradigm",slug:"electrically-small-microstrip-antennas-targeting-miniaturized-satellites-the-cubesat-paradigm",signatures:"Constantine Kakoyiannis and Philip Constantinou",authors:[{id:"19252",title:"Dr.Ing.",name:"Constantine",middleName:"G.",surname:"Kakoyiannis",fullName:"Constantine Kakoyiannis",slug:"constantine-kakoyiannis"},{id:"21863",title:"Prof.",name:"Philip",middleName:null,surname:"Constantinou",fullName:"Philip Constantinou",slug:"philip-constantinou"}]},{id:"14718",title:"Circularly Polarized Microstrip Antennas with Proximity Coupled Feed for Circularly Polarized Synthetic Aperture Radar",slug:"circularly-polarized-microstrip-antennas-with-proximity-coupled-feed-for-circularly-polarized-synthe",signatures:"Merna Baharuddin and Josaphat Tetuko Sri Sumantyo",authors:[{id:"18521",title:"PhD.",name:"Merna",middleName:null,surname:"Baharuddin",fullName:"Merna Baharuddin",slug:"merna-baharuddin"},{id:"21582",title:"Prof.",name:"Josaphat",middleName:null,surname:"Tetuko Sri Sumantyo",fullName:"Josaphat Tetuko Sri Sumantyo",slug:"josaphat-tetuko-sri-sumantyo"}]},{id:"14719",title:"Circularly Polarized Slotted/Slit-Microstrip Patch Antennas",slug:"circularly-polarized-slotted-slit-microstrip-patch-antennas",signatures:"Nasimuddin, Zhi-Ning Chen and Xianming Qing",authors:[{id:"21459",title:"Dr.",name:"N",middleName:null,surname:"Nasimuddin",fullName:"N Nasimuddin",slug:"n-nasimuddin"}]},{id:"14720",title:"Microstrip Antenna Arrays",slug:"microstrip-antenna-arrays",signatures:"Albert Sabban",authors:[{id:"16889",title:"Dr.",name:"Albert",middleName:null,surname:"Sabban",fullName:"Albert Sabban",slug:"albert-sabban"}]},{id:"14721",title:"Microstrip Antennas for Indoor Wireless Dynamic Environments",slug:"microstrip-antennas-for-indoor-wireless-dynamic-environments",signatures:"Mohamed Elhefnawy and Widad Ismail",authors:[{id:"17004",title:"Dr.",name:"Widad",middleName:null,surname:"Ismail",fullName:"Widad Ismail",slug:"widad-ismail"},{id:"17005",title:"Dr.",name:"Mohamed",middleName:null,surname:"Elhefnawy",fullName:"Mohamed Elhefnawy",slug:"mohamed-elhefnawy"}]},{id:"14722",title:"DBDP SAR Microstrip Array Technology",slug:"dbdp-sar-microstrip-array-technology",signatures:"Shun-Shi Zhong",authors:[{id:"4123",title:"Prof.",name:"Shun-Shi",middleName:null,surname:"Zhong",fullName:"Shun-Shi Zhong",slug:"shun-shi-zhong"}]},{id:"14723",title:"Microwave Properties of Dielectric Materials",slug:"microwave-properties-of-dielectric-materials",signatures:"JS Mandeep and Loke Ngai Kin",authors:[{id:"21035",title:"Prof.",name:"Mandeep",middleName:null,surname:"Singh Jit",fullName:"Mandeep Singh Jit",slug:"mandeep-singh-jit"},{id:"135784",title:"Prof.",name:"Ngai Kin",middleName:null,surname:"Loke",fullName:"Ngai Kin Loke",slug:"ngai-kin-loke"}]},{id:"14724",title:"Hybrid Microstrip Antennas",slug:"hybrid-microstrip-antennas",signatures:"Alexandre Perron, Tayeb A. Denidni and Abdel R. Sebak",authors:[{id:"11473",title:"Prof.",name:"Tayeb A.",middleName:null,surname:"Denidni",fullName:"Tayeb A. Denidni",slug:"tayeb-a.-denidni"},{id:"21901",title:"Prof.",name:"Alexandre",middleName:null,surname:"Perron",fullName:"Alexandre Perron",slug:"alexandre-perron"},{id:"21902",title:"Prof.",name:"Abdel R.",middleName:null,surname:"Sebak",fullName:"Abdel R. Sebak",slug:"abdel-r.-sebak"}]},{id:"14725",title:"Integration of 60-GHz Microstrip Antennas with CMOS Chip",slug:"integration-of-60-ghz-microstrip-antennas-with-cmos-chip",signatures:"Gordana Klaric Felic and Efstratios Skafidas",authors:[{id:"18389",title:"Prof.",name:"Gordana Klaric",middleName:null,surname:"Felic",fullName:"Gordana Klaric Felic",slug:"gordana-klaric-felic"},{id:"21215",title:"PhD.",name:"Efstratios",middleName:null,surname:"Skafidas",fullName:"Efstratios Skafidas",slug:"efstratios-skafidas"}]},{id:"14726",title:"A Practical Guide to 3D Electromagnetic Software Tools",slug:"a-practical-guide-to-3d-electromagnetic-software-tools",signatures:"Guy A. E. Vandenbosch and Alexander Vasylchenko",authors:[{id:"18667",title:"PhD.",name:"Alexander",middleName:null,surname:"Vasylchenko",fullName:"Alexander Vasylchenko",slug:"alexander-vasylchenko"},{id:"20908",title:"Prof.",name:"Guy A. E.",middleName:null,surname:"Vandenbosch",fullName:"Guy A. E. Vandenbosch",slug:"guy-a.-e.-vandenbosch"}]}]}]},onlineFirst:{chapter:{type:"chapter",id:"67998",title:"Electric Transmission Network Expansion Planning with the Metaheuristic Variable Neighbourhood Search",doi:"10.5772/intechopen.87071",slug:"electric-transmission-network-expansion-planning-with-the-metaheuristic-variable-neighbourhood-searc",body:'\n
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1. Introduction
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Due to consumption growth of electrical power, the need of increasing the existing transmission network power flow capacity is evident. This expansion can be a dynamic or static performance. The static long-term power transmission network expansion planning (TNEP) problem consists of determining the minimum cost planning which specifies the number and the locations of transmission lines to meet a forecasted demand while satisfying the balance between generation and load and other operational constraints [1]. Transmission investments are very capital intensive and have long useful lives, so transmission investment decisions have a long-standing impact on the power system as a whole; therefore TNEP has become an important component of power system planning, and its solution is used to guide future investment in transmission equipment.
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The pioneering work on transmission expansion planning is reported in [2], and since then TNEP literature has been vast and reports that there are usually considered various solution methods that depend on the mathematical model formulation [3]. A state of the art, which was obtained from the review of the most interesting models found in the international technical literature, is presented in [4]. In [5] TNEP is reviewed from different aspects such as modelling, solving methods, reliability, distributed generation, electricity market, uncertainties, line congestion and reactive power planning. A critical review focusing on its most recent developments and a taxonomy of modelling decisions and solution to TNEP are presented in [6].
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The convenient mathematical modelling to indicate the appropriate operation would be the representation of the problem by mathematical relationships of the AC load flow, typically used for the electric system operation analysis [1]. However, this modelling is more difficult to be used in an efficient way in transmission network planning, due to its non-convex and nonlinear nature. Consequently, the mathematical modelling in its most accurate representation is the direct current (DC) model, which considers Kirchhoff’s voltage (KVL) and current (KCL) laws just for balance and active power flow. In this case, the resulting problem is a nonlinear mixed-integer programming with high complexity for large systems, presenting combinatorial explosion of the number of alternative solutions, with extra difficulty of presenting many local optima, which most of the time are of poor quality [3].
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A more simplified modelling is the so-called transportation model (TM) which just enforces the KLC at all existent nodes [2]. In this case the resulting problem is an integer linear programming problem which is normally easier to solve than the DC model although it maintains the combinatorial characteristic of the original problem [3].
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It is still possible to consider hybrid models which combine characteristics of the DC model and the transportation model. In this model it is assumed that KCL constraints are satisfied for all nodes of the network, whereas the constraint which represents Ohm’s law (and indirectly KVL) is satisfied only by the existing circuits (and not necessarily by the added circuits) [3].
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Technical literature related to the TNEP proposes many solution methods that can be classified into mathematical optimization, heuristic and metaheuristic approaches [7]. Techniques such as dynamic programming [8], linear programming [2], nonlinear programming [9], mixed-integer programming [10], branch and bound [11], hierarchical decomposition [12] and Benders decomposition [13] have been used and are categorized as mathematical-based approaches. But these techniques demand large computing time due to the dimensionality curse of this kind of problem. Heuristic methods emerged as an alternative to classical optimization methods, and their use has been very attractive since they were able to find good feasible solutions demanding less computational effort.
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Some heuristic approaches have been proposed using constructive heuristic algorithms (CHA) [10, 14, 15, 16] and the forward-backward approach [17]. Metaheuristic methods emerged as an alternative to the two previous approaches, producing high-quality solutions with moderate computing time. Genetic algorithms [18, 19], greedy randomized adaptive search procedure [13], tabu search [20, 21], simulated annealing [20, 22], GRASP [23], scatter search [24] and grey wolf optimization algorithm [25] have been used to solve the TNEP problem, among other metaheuristic optimization techniques. It is important to point out that they cannot guarantee the global optimal solution to the TNEP problem.
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A varied bibliography regarding the theory and application of metaheuristics can be found in [26, 27]. Other applications of metaheuristics appear in [28].
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Considering that exact methods of optimization to TNEP are not efficient to big data problems, this paper presents a novel metaheuristic method that considers the so-called variable neighbourhood search (VNS) to solve the TNEP problem considering DC model. The VNS metaheuristic was presented in the middle of the 1990s, by Mladenovic and Hansen [29], and represents a significantly different proposal compared to other metaheuristics. The fundamental idea of the VNS algorithm is based on a basic principle: to explore the space of solutions by systematic changes of neighbourhood structures during the search process. Thus, the transition through the search space of the problem is always accomplished with an improvement of the objective function, and, therefore, the transition is not allowed for a solution of worse quality as occurs with most of the metaheuristics [29].
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The VNS algorithm was used with success in the optimization of several problems of operational research [26, 27, 29, 30], but it is still insignificant in the optimization of problems related to the operation and the planning of electric power systems. The VNS was used to TNEP considering transportation model in [31, 32].
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This paper is organized as follows: Initially the mathematical model for TNEP problem and the VNS metaheuristic are presented. After, the developed VNS algorithm to solve the TNEP problem is described. Later, obtained results are presented and commented. Finally, conclusions are drawn.
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2. Mathematical model of TNEP
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The mathematical formulation of the TNEP for the DC model is given by Eqs. (1)–(8) and performs as a nonlinear mixed-integer programming problem [3]:
where \n\nv\n\n is the total investment value for a predefined horizon; \n\n\nc\nij\n\n\n is the cost of a circuit or facility that can be added in the branch \n\n\ni\nj\n\n\n; \n\n\nn\nij\n\n\n is the number of circuits added during the optimization process; \n\n\nn\nij\n0\n\n\n is the number of existing circuits in the initial topology; \n\n\nγ\nij\n\n\n is the susceptance of the branch \n\n\ni\nj\n\n\n; \n\n\nθ\ni\n\n\n is phase angle at the bus \n\ni\n\n; \n\nF\n\n\nis the vector of power flow with components \n\n\nf\nij\n\n\n; \n\n\n\nf\n¯\n\nij\n\n\n is the transmission capacity of a circuit through branch \n\n\ni\nj\n\n\n; \n\nA\n\n is the transposed incidence branch-node matrix of the power system; \n\nG\n\n is a vector with elements \n\n\ng\nk\n\n\n (power generation at bus \n\nk\n\n) with maximum values \n\n\n\ng\n¯\n\nk\n\n\n; \n\n\n\nn\n¯\n\nij\n\n\n is the maximum number of circuits that can be added to the branch \n\n\ni\nj\n\n\n; \n\nΩ\n\n is the set of all branches where it is possible to add new circuits.
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Eq. (1) that contains the sum of the investments costs is the objective function. The KCL is framed in Eq. (2), and the Ohm’s law is expressed in Eq. (3) which implicitly takes into consideration Kirchhoff’s voltage law (KVL). Inequalities Eq. (4) represent capacity constraints for transmission lines, whereas the absolute value is necessary since power can flow in both directions. Other constraints Eqs. (6)–(8) represent operational limits of the generators, maximum limit for the addition of circuits per branch and integrality demand of the variables \n\n\nn\nij\n\n\n, respectively.
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The model Eqs. (1)–(8) cannot be solved by using traditional algorithms, and there is no efficient method for solving these kinds of problems directly. Therefore, metaheuristics become suitable optimization tools for finding optimal and suboptimal solutions for the TNEP problem when it is considered complex power systems (big instances).
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A more simplified model called the transport model can be considered, which contemplates only Kirchhoff’s current law and could be obtained by relaxing the nonlinear constraint Eq. (3) of the DC model described above [3]. In this case, the resulting model is an integer linear programming problem. Even though it is linear, it is still very difficult to find the optimal solution for large and complex systems. The transport model was the first systematic proposal of mathematical modelling used with great success in the problem of planning of transmission systems. The model was proposed by Garver [2] and has represented the beginning of systematic research in the area of transmission system planning.
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Another model that has been considered for the PPEST is the linear hybrid model (LHM) which combines characteristics of the DC model and the transport model. This model, in a simpler formulation, preserves the linear properties of the transport model, considering Kirchhoff’s current law in all nodes of the network and KVL only in the circuits in the base network (not necessarily in the circuits that will be added) [3, 10]. The LHM is framed by Eqs. (9)–(17):
where \n\n\nA\n0\n\n\n is the transposed incidence branch-node matrix of the base topology in previews iterations of the algorithm system; \n\n\nF\n0\n\n\n is the vector of base power flow with components \n\n\nf\nij\n0\n\n\n; \n\n\nn\nij\n0\n\n\n is the circuits added during the iterative process to the base case; \n\n\nΩ\n0\n\n\n is the set of all the circuits added during the iterative process and all of the prime circuits of the base case.
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The LHM was originally proposed in [10] whose authors present a mathematical modelling Eqs. (9)–(17) which specifies that the portion of the electric system corresponding to the circuits existing in the base configuration must satisfy the two Kirchhoff’s laws and the other corresponding part from new circuits must satisfy only Kirchhoff’s current law.
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The LHM Eqs. (9)–(17) will be considered as a sensitivity indicator to the proposed heuristic algorithm.
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3. Metaheuristic VNS
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A metaheuristic is a search strategy that orchestrates an interaction between local improvement procedures and higher local strategies to create a process capable of escaping from local optima and performing a robust optimization method for complex problems. This search is performed by means of transitions in the search space from an initial solution or a set of initial solution. In this context, the main difference among the diverse metaheuristic techniques is the strategy used to carry out the transitions within the search space. VNS is a metaheuristic that systematically exploits the idea of neighbourhood change to find local-optimal solutions and to leave those local optima. In that fundamental aspect, VNS is significantly different from other metaheuristics. Most metaheuristics accept the degradation of the current solution as a strategy to leave a local-optimal solution. The VNS algorithm does not accept this possibility [26].
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The VNS algorithm changes the neighbourhood as a way of leaving local-optimal solutions. During this process, the current solution is also the incumbent, which does not happen with other metaheuristics. Thus, it is possible to state that the VNS algorithm performs a set of transitions in the search space of a problem and at each step this transition is performed for the new incumbent. If the process finds a local optimum, then the VNS algorithm changes the neighbourhood in order to leave from that local optimum and to achieve the new incumbent. As a consequence of this strategy, if the VNS algorithm finds the global optimum, the search stops at that point, eliminating any chance of leaving it. This behaviour does not occur with other metaheuristics.
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The strategy of the VNS algorithm is inspired by three important facts [29]:
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Fact 1—A minimum with regard to one neighbourhood structure is not necessary for another.
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Fact 2—A global minimum is a local minimum with regard to all possible neighbourhood structures.
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Fact 3—For many problems, a local minimum with regard to one or several neighbourhoods is relatively close to each other.
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The latter is particularly important in the formulation of the VNS algorithm. This empirical fact implies that a local-optimal solution often provides important information regarding the global one, especially if the local-optimal solution presents excellent quality. It is also an empirical fact that local-optimal solutions are generally concentrated in specific regions of the search space. If local-optimal solutions were to be uniformly distributed in the search space, all metaheuristics would become inefficient. Consequently, if a local optimum is found in the same region where the global optimum is, then the VNS metaheuristic has better chances of finding this global optimum. On the other hand, if the global optimum pertains to another region, then the only possibility to find it is to implement a diversification process. For this reason, equilibrium between intensification and diversification during the search process can be important in a metaheuristic.
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There is another important aspect related to the quality of the local optimum that should be part of the implementation logic of a VNS algorithm. A local optimum with a better-quality objective function is not necessarily more suitable for trying to find the global optimum. Let \n\n\nx\na\n\n\n and \n\n\nx\nb\n\n\n be two local-optimal solutions with \n\nf\n\n\nx\na\n\n\n<\nf\n\n\nx\nb\n\n\n\n for the minimization problem. Considering the traditional analysis, it can be concluded that \n\n\nx\na\n\n\n is a local optimum with better quality than \n\n\nx\nb\n\n\n.
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If these solutions are to be used for initiating (or reinitiating) the search process, then it can be affirmed that the solution presenting internal characteristics closer to those of the global optimum is the most suitable for initiating (or reinitiating) the search and, consequently, solution should not necessarily be chosen.
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Thus, for instance, considering the TNEP problem, the local-optimal solution with the largest number of \n\n\nn\nij\n\n\n elements equal to the optimal solution is the most appropriate for initiating (or reinitiating) the search. It is evident that in normal conditions, the optimal solution is unknown. However, there are some problems where the optimal solution is known, and there are also various heuristic algorithms to find local-optimal solutions for this problem.
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In this way, the previous observation can be used to identify the heuristic algorithm that produces best-quality local-optimal solutions for initiating the search using the VNS algorithm. This type of behaviour occurs in the TNEP problem where for some instances (power systems) optimal solutions are known and various constructive heuristic algorithms used to find excellent local-optimal solutions are available. Thus, the best constructive heuristic algorithm to be incorporated into the solution structure of a VNS algorithm can be identified.
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Let \n\n\nN\nk\n\n\n, \n\nk\n=\n1\n,\n…\n,\n\nk\nmax\n\n\n be a finite set of preselected neighbourhood structures, and let \n\n\nN\nk\n\n\nx\n\n\n be a set of solutions or neighbours in the \n\nk\n\nth neighbourhood of \n\nx\n\n.
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An optimal solution \n\n\nx\nopt\n\n\n (or global minimum) is a solution where the minimum of Eqs. (9)–(17) is achieved.
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A solution \n\n\nx\n′\n\n\n is a local minimum of Eqs. (1)–(8) with regard to \n\n\nN\nk\n\n\nx\n\n\n, if there is no solution \n\n\nx\n′\n\n∈\n\nN\nk\n\n\nx\n\n⊆\nX\n,\n\n such that \n\nf\n\n\nx\n\n′\n′\n\n\n\n<\nf\n\n\nx\n′\n\n\n\n.
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Thus, the idea is to define a set of neighbourhood structures that can be used in a deterministic, random or both deterministic and random manners. These different forms of using the neighbourhood structure lead to VNS algorithms with different performances.
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There are various proposals of VNS algorithms that can be used independently or in an integrated manner forming more complex VNS structures. The simplest form of a VNS algorithm is the variable neighbourhood descent (VND). The VND algorithm is based on previously mentioned Fact 1, i.e. the local minimum for a given move is not necessarily the local minimum for another type of move [29]. In this way, the local optimum \n\n\nx\n′\n\n\n in the neighbourhood \n\n\nN\n1\n\n\nx\n\n\n is not necessarily equal to the local optimum \n\n\nx\n\n′\n′\n\n\n\n of \n\n\nx\n′\n\n\n to the neighbourhood \n\n\nN\n2\n\n\nx\n\n\n.
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The VND algorithm takes on the form shown in Figure 1.
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Figure 1.
VND algorithm [33].
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This algorithm can be integrated into a more complex structure of the VNS algorithm.
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For example, the sept (a) in Figure 1 could be replaced by randomly generating a solution neighbour \n\n\nx\n′\n\n\n of \n\nx\n(\n\nx\n′\n\n∈\n\nN\nk\n\n\nx\n\n\n); and the resulting algorithm is called the reduced variable neighbourhood search (RVNS). In the RVNS, usually, the neighbourhoods will be nested, i.e. each one contains the previous. Then a point is chosen at random in the first neighbourhood. If its value is better than that of the incumbent (i.e. \n\nf\n\n\nx\n′\n\n\n<\nf\n\nx\n\n\n), the search is recentred there (\n\n\nx\n′\n\n\n ← \n\nx\n\n). Otherwise, one proceeds to the next neighbourhood. After all neighbourhoods have been considered, one begins again with the first, until a stopping condition is met.
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The RVNS algorithm chooses neighbours more dynamically by selecting those from all neighbourhood structures (diversification) and prioritizing the first neighbourhood structure (intensification) during the initial stages of the search. Nevertheless, an important component of the RVNS structure is its capacity for finding new promising regions from a local optimum. The RVNS algorithm can also be used independently or be integrated into a more complex structure of the VNS algorithm.
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More efficient VNS algorithms can be formulated by integrating those characteristics of the VND algorithm that allow local quality optima to be found and those of the RVNS algorithm that allow new promising regions from a local optimum to be found. Thus, by merging those characteristics, two types of VNS algorithms that generally exhibit excellent performance can be formulated. These algorithms are called the basic variable neighbourhood search (BVNS) and the general variable neighbourhood search (GVNS).
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The BVNS algorithm combines a local search with systematic changes of neighbourhood around the local optimum found in [33]. The structure of the BVNS algorithm is presented in Figure 2.
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Figure 2.
BVNS framework [33].
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The logical procedure adopted by the BVNS is very interesting. Firstly, \n\nk\n\n\nneighbourhood structures should be chosen. The optimization process is initiated from a solution \n\nx\n\n\nand the corresponding neighbourhood \n\n\n\nN\n1\n\n\nx\n\n\n. Then, a neighbour \n\nx\n′\n\n of \n\nx\n\n\nin \n\n\nN\n1\n\n\nx\n\n\n is randomly selected. From \n\nx\n′\n\n, a local search process to find the local optimum \n\nx\n′\n′\n\n is started.
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In this context, three cases may occur:
If \n\nx\n′\n′\n\n it is equal to \n\nx\n′\n\n, one already was the local optimum of the valley and, consequently, a change of neighbourhood level should be performed (\n\n\nN\n2\n\n\nx\n\n\n in this case).
If \n\nx\n′\n′\n\n is worse than \n\n\nx\n′\n\n,\n\n then the local optimum with less quality than the incumbent \n\nx\n\n was found, and a change of neighbourhood should also be carried out.
If \n\nx\n′\n′\n\n is better than \n\nx\n′\n\n, it means that a better solution than the incumbent was found, and, consequently, the incumbent should be updated; the search should be reinitiated from the new incumbent while remaining in the neighbourhood \n\n\nN\n1\n\n.\n\n
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Whenever the local search finds a new incumbent, at any iteration of the process, the neighbourhood \n\n\nN\n1\n\n\nx\n\n\n should be considered again. Also, whenever the local search finds an equal or worse quality solution than the incumbent, a change towards a more complex neighbourhood should be performed. This strategy and the random choice of the incumbent \n\nx\n’\n\n neighbour avoid cycling and allow local optima which are distant from the current incumbent to be found.
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The local search of the BVNS algorithm can be any heuristic strategy. Nonetheless, the local search can also use a strategy of the VNS algorithm. Therefore, the BVNS algorithm can be transformed into a more general algorithm called general variable neighbourhood search (GVNS). The GVNS algorithm is obtained through the generalization of the BVNS algorithm by simply using a VND algorithm as a local search and using a RVNS algorithm to improve the initial solution required to begin the search.
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All observations made for the BVNS algorithm remain valid for the GVNS algorithm. As mentioned previously, the fundamental change corresponds to the improvement stage of the initial solution using an RVNS algorithm and a VND algorithm for the local search stage.
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Since the VNS algorithm can be implemented in various ways, a family of VNS algorithms can also be implemented. In [26, 30, 33] diverse types of VNS algorithms are analysed. In this work, only one of these algorithms is presented. There are other more complex algorithms or structures based on the logic of the VNS algorithm that are out of the scope of this work. Those algorithms can be found in [30, 33].
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4. Modified VNS for TNEP
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In this section the application of our proposed VNS to the TNEP will be described. The GVNS described in Figure 3 will be used considering the following steps that will be explained in detail in sequence:
Step 1—Initial solution: Considering a heuristic algorithm to determine an initial solution.
Step 2—Definition of neighbourhoods: Characterization of each neighbourhood and determination of their elements.
Step 3—Improvement: Improve the initial solution by using a RVNS algorithm.
Step 4—Local search: Apply some local search to determine the best configuration for each current solution neighbourhood.
Step 1: Initial solution
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Figure 3.
GVNS framework [33].
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To determine a DC initial solution to TNEP, the constructive heuristic algorithm (CHA) presented by Villasana-Garver-Salon (VGS) [10] is considered. This algorithm iteratively chooses a new circuit to be added to the system considering a step-by-set procedure that uses a sensitivity index (given in Eq. (18)) that plays a key role in the CHA. The iteratively process continues until a feasible solution is achieved; that means that there is no need for new circuit additions:
Generally, for large and complex systems, the derived solutions are local-optimal [10]. The VGS can be summarized by the following steps:
VGS1: Take a base topology as a current solution, and resolve the HML Eqs. (10)–(17) considering that all of the circuits of the current solution must follow both Kirchhoff’s laws.
VGS2: Solve LP for the HML using the current solution. If the LP solution indicates that the system is adequately operating with the new additions and \n\nv\n=\n0\n\n, then stop. A new solution for the DC model was found. Go to step 4.
VGS3: Identify the most attractive circuit considering the sensitivity in Eq. (18). Update the current solution with the chosen circuit, update \n\n\nn\nij\n0\n\n\n and \n\n\nΩ\n0\n\n\n, and go to step 2.
\n
All of the added circuits represent the solution of the CHA. It can be noted that although the VGS uses a hybrid linear model to identify the best circuit for addition in an iterative process, it complies with both of Kirchhoff’s laws after adding a new circuit; thus, the final solution is also feasible in DC.
\n
Example 1: considering Graver’s system [34] that includes six transmission lines and six buses with a 760-MW demand for base topology, which is shown in Figure 4a, after has applied the VGS it gave the topology in Figure 4b, with \n\nv\n=\n130.000\n\n m.u.
Step 2: Definition of neighbourhoods
\n
Figure 4.
Base topology and VGS solution for Graver’s system. (a) Base topology and (b) Initial solution by VGS.
\n
Given solution \n\nx\n,\n\n the structures of neighbourhood within the solution space can be defined by Eq. (19):
where \n\nd\n\nx\n\nx\n′\n\n\n=\nk\n\n is the quantity of branches with a different number of added circuits in the solutions \n\nx\n\n and \n\nx\n′\n\n.
\n
For example, given solutions \n\nx\n\n, \n\nx\n’\n\n and \n\nx\n”\n\n from Figure 5a–c, respectively, which are coded in Figure 6, \n\nd\n\nx\n\nx\n′\n\n\n=\n1\n\n, and \n\nd\n\nx\n\nx\n\n′\n′\n\n\n\n=\n2\n\n. So, solution \n\nx\n′\n\n is a neighbour of \n\nx\n\n in \n\n\nN\n1\n\n\nx\n\n\n, and solution \n\nx\n′\n′\n\n is a neighbour of \n\nx\n\n in \n\n\nN\n2\n\n\nx\n\n\n.
\n
Figure 5.
Neighbourhood characterization.
\n
Figure 6.
x, x′ and x′′ neighbours codification.
\n
Neighbour \n\nx\n′\n\nis obtained from \n\nx\n\n by adding a circuit in branch 8 (buses 3–6), whereas the neighbour \n\nx\n′\n′\n\n was obtained from \n\nx\n\n by adding one circuit in branch 7 (buses 3–5) and removing one circuit in branch 9 (buses 4–6). In the same way, the neighbours in the other \n\nk\n\n neighbourhoods can be obtained.
Step 3: Improvement of the initial solution
\n
Considering \n\n\nk\nmax\n\n=\n5\n\n and the initial solution obtained in step 1, a local improvement search using a GVNS described in Figure 3 is applied considering the HLM Eqs. (9)–(17).
\n
In \n\n\nN\n1\n\n\nx\n\n\n, sort all added circuits in cost-decreasing order, remove the circuit having the maximum cost, and verify the operation using the HLM model. If such removal keeps a feasible solution which indicates that the system is in adequate operation condition (i.e. \n\nv\n=\n0\n\n after HML solution), remove that circuit; otherwise, keep the circuit. Repeat the process of simulating circuit removal until all of the added circuits have been tested.
\n
At the end of the process, all the added circuits that were not removed represent the improved solution.
\n
As for the remaining neighbourhoods, the cost variations due to changes (cost difference between entering and leaving circuits) are calculated, and only the changes that exhibit negative variation are simulated (the HLM is solved). If the simulation points out a feasible configuration, then it is a candidate to be used by updating the current configuration. If the new configuration is unfeasible, then the simulation is cancelled.
\n
It is important to elucidate that the movement only be carried out if the new configuration is better than the incumbent and that in this step the procedure only accepts movements that lead to feasible solutions.
\n
The stop criterion corresponds to the maximum number of solved HLM.
Step 4: Local search
\n
The local search is based in VND described in Figure 1.
\n
\n
\n
5. Results
\n
To illustrate the effectiveness of the proposed method, three problems are considered: the Garver 6-bus, the IEEE 24-bus and the Brazilian Southern 46-bus systems.
\n
Full data can be found in [34, 35, 36], respectively. Planning could be done with (r) or without (w) generation rescheduling, resulting in these following cases that have been widely used to validate results of new methods [2, 10, 15, 16, 20, 34]; Da [21, 22, 23, 24, 31, 32, 35, 36]:
Case 1w: Garver 6-bus system without rescheduling
Case 1r: Garver 6-bus system with rescheduling
Case 2w: IEEE 24-bus system without rescheduling
Case 2r: IEEE 24-bus system with rescheduling
Case 3w: Brazilian Southern 46-bus system without rescheduling
Case 3r: Brazilian Southern 46-bus system with rescheduling
\n
The Brazilian Southern is a real referred system originally formed by 46 buses and 66 circuits in the base topology, 79 candidate paths and 6.880 MW as expected demand [35].
\n
For reducing the size of the considered neighbourhoods, only those added circuits operating below 70% of their capacity were considered to be candidate circuits for removal.
\n
Table 1 shows the results. The proposed method was more efficient than the methods shown in [15, 20], since it requires less number of linear programing resolutions.
In this paper an efficient new method based on variable neighbourhood search has been proposed for transmission networking problem planning considering the DC model whose mathematical formulation is nonlinear and mixed integer. The TNEP is a multimodal problem of high complexity for medium and large systems and cannot be solved by exact algorithms in reasonable computational times.
\n
The proposed method systematically exploits the idea of neighbourhood change to find local-optimal solutions and to leave those local optima. It was observed that the definition of neighbourhood structures plays an important role to the convergence of the VNS algorithm applied to TNEP.
\n
The proposed method was tested in the Garver 6-bus, in the IEEE 24-bus and in the Brazilian Southern 46-bus systems, and the results got more chance of finding better solutions than mathematical optimization techniques and find local-optimal solution requiring fewer solved linear problems.
\n
As further research directions, new strategies for reducing the size of the neighbourhood such as using adjacency lists to avoid adding new lines in isolated circuits and different kinds of structure neighbourhoods could be developed.
\n
\n\n',keywords:"transmission network expansion planning, variable neighbourhood search algorithm, metaheuristic algorithm, power system planning, combinatorial optimization",chapterPDFUrl:"https://cdn.intechopen.com/pdfs/67998.pdf",chapterXML:"https://mts.intechopen.com/source/xml/67998.xml",downloadPdfUrl:"/chapter/pdf-download/67998",previewPdfUrl:"/chapter/pdf-preview/67998",totalDownloads:128,totalViews:0,totalCrossrefCites:0,dateSubmitted:"June 14th 2018",dateReviewed:"May 27th 2019",datePrePublished:"July 5th 2019",datePublished:null,readingETA:"0",abstract:"This paper presents a new method to solve the static long-term power transmission network expansion planning (TNEP) problem that uses the metaheuristic variable neighbourhood search (VNS). The TNEP is a large-scale, complex mixed-integer nonlinear programming problem that consists of determining the optimum expansion in the network to meet a forecasted demand. VNS changes structure neighbourhood within a local algorithm and makes the choices of implementation that integrate intensification and/or diversification strategies during the search process. The initial solution is obtained by a heuristic nonlinear mixed integer which takes two Kirchhoff’s laws (transportation and the DC models have been used). Several tests are performed on Graver’s 6-bus, IEEE 24-bus and Southern Brazilian systems displaying the applicability of the proposed method, and results show that the proposed method has a significant performance in comparison with some studies addressed in common literature.",reviewType:"peer-reviewed",bibtexUrl:"/chapter/bibtex/67998",risUrl:"/chapter/ris/67998",signatures:"Silvia Lopes de Sena Taglialenha and Rubén Augusto Romero Lázaro",book:{id:"7607",title:"Artificial Neural Networks",subtitle:null,fullTitle:"Artificial Neural Networks",slug:null,publishedDate:null,bookSignature:"Dr. Ali Sadollah and Dr. Carlos M. Manuel Travieso-Gonzalez",coverURL:"https://cdn.intechopen.com/books/images_new/7607.jpg",licenceType:"CC BY 3.0",editedByType:null,editors:[{id:"147215",title:"Dr.",name:"Ali",middleName:null,surname:"Sadollah",slug:"ali-sadollah",fullName:"Ali Sadollah"}],productType:{id:"1",title:"Edited Volume",chapterContentType:"chapter",authoredCaption:"Edited by"}},authors:null,sections:[{id:"sec_1",title:"1. Introduction",level:"1"},{id:"sec_2",title:"2. Mathematical model of TNEP",level:"1"},{id:"sec_3",title:"3. Metaheuristic VNS",level:"1"},{id:"sec_4",title:"4. Modified VNS for TNEP",level:"1"},{id:"sec_5",title:"5. Results",level:"1"},{id:"sec_6",title:"6. Conclusions",level:"1"}],chapterReferences:[{id:"B1",body:'Sullivan RL. Power System Planning. New York: McGraw-Hil; 1977\n'},{id:"B2",body:'Garver LL. Transmission network estimation using linear programming. IEEE Transaction Apparatus Systems. 1970;89:1688-1697\n'},{id:"B3",body:'Romero R, Monticelli A, Garcia A, Haffner S. Test systems and mathematical models for transmission network expansion planning. IEE Proceedings Generation, Transmission and Distribution. 2002;149(1):27-36\n'},{id:"B4",body:'Latorre G, Cruz RD, Areiza JM, Villegas A. Classification of publications and models on transmission expansion planning. IEEE Transactions on Power Systems. 2003;18(2):938-946\n'},{id:"B5",body:'Hemmati R, Hooshmand RA, Khodabakhshian A. State-of-the-art of transmission expansion planning: Comprehensive review. Renewable and Sustainable Energy Reviews. 2013;23:312-319. DOI: 10.1016/j.rser.2013.03.015\n'},{id:"B6",body:'Lumbreras S, Ramos A. The new challenges to transmission expansion planning. Survey of recent practice and literature review. Electric Power Systems Research. 2016;134:19-29\n'},{id:"B7",body:'Lee CW, Ng SKK, Zhong J, Wu FF. Transmission Expansion Planning From Past to Future. In: IEEE PES Power Systems Conference and Exposition; Atlanta, GA; 2006. pp. 257-265\n'},{id:"B8",body:'Dusonchet YP, El-Abiad A. Transmission planning using discrete dynamic optimizing. IEEE Transactions on Power Apparatus and Systems. 1973;PAS-92(4):1358-1137. DOI: 10.1109/TPAS.1973.293543\n'},{id:"B9",body:'Al-Hamouz ZM, Al-Faraj AS. Transmission expansion planning using nonlinear programming. In: IEEE/PES Transmission and Distribution Conference and Exhibition; Vol. 1. Yokohama, Japan: IEEE; 2002. pp. 50-55. DOI: 10.1109/TDC.2002.1178259\n'},{id:"B10",body:'Villasana R, Garver LL, Salon SJ. Transmission network planning using linear programming. IEEE Transactions on Power Systems. 1985;104:349-356\n'},{id:"B11",body:'Haffner S, Monticelli A, Garcia A, Mantovani J, Romero R. Branch and bound algorithm for transmission system expansion planning using a transportation model. IEE Proceedings - Generation, Transmission and Distribution, V. 2000;147(3):149-156\n'},{id:"B12",body:'Romero R, Monticelli A. A hierarchical decomposition approach for transmission network expansion planning. IEEE Transactions on Power Systems. 1994;9:373-380\n'},{id:"B13",body:'Binato S, Pereira MVF, Granville S. A new benders decomposition approach to solve power transmission network design problems. IEEE Transactions on Power Apparatus and Systems. 2001;16:235-240\n'},{id:"B14",body:'Romero R, Rocha C, Mantovani JRS, Sanchez IG. Constructive heuristic algorithm for the DC model in network transmission expansion planning. IEE Proceedings-Generation, Transmission and Distribution. 2005;152(2):277-282\n'},{id:"B15",body:'Romero R, Rider M, Silva I. A metaheuristic to solve the transmission expansion planning. IEEE Transactions on Power Systems. 2007;22:2289-2291\n'},{id:"B16",body:'Da Silva EL, Gil HA, Areiza JM. Transmission network expansion planning under an improved genetic algorithm. IEEE Transactions on Power Systems. 2000;15(4):1168-1117\n'},{id:"B17",body:'Seifi H, Sepasian MS, Haghighat H, Foroud AA, Yousefi GR, Rae S. Multi-voltage approach to long-term network expansion planning. IET Generation Transmission and Distribution. 2007;1:9. DOI: 10.1049/iet-gtd:20070092\n'},{id:"B18",body:'Gallego RA, Monticelli A, Romero R. Transmission expansion planning by extended genetic algorithm. IEE Proceedings - Generation, Transmission and Distribution. 1998:145(3):329-335\n'},{id:"B19",body:'da SilvaEL, Gil HA, Areiza JM. Transmission network expansion planning under an improved genetic algorithm. IEEE Transmission Power Systems. 2000;15:1168-1175\n'},{id:"B20",body:'Gallego RA, Monticelli A, Romero R. Comparative studies of non-convex optimization methods for transmission network expansion planning. IEEE Transactions on Power Systems. 1998;13(3):822-828\n'},{id:"B21",body:'Da Silva EL, Areiza JM, Oliveira GC, Binato S. Transmission network expansion planning under a tabu search approach. IEEE Transactions on Power Systems. 2001;16(1):62-68\n'},{id:"B22",body:'Gallego RA, Alves AB, Monticelli A, Romero R. Parallel simulated annealing applied to long term transmission expansion planning. IEEE Transactions on Power Systems. 1997;1(12):181-187\n'},{id:"B23",body:'Faria HJ, Binato S, Resende MGC, Falcão DM. Power transmission network design by greedy randomized adaptive path relinking. IEEE Transactions on Power Systems. 2005;20(1):43-49\n'},{id:"B24",body:'Mori H, Shimomugi K. Network expansion planning with scatter search. In: IEEE International Conference on Systems, Man and Cybernetics. ISIC; 2007. pp. 3749-3754\n'},{id:"B25",body:'Khandelwal A, Bhargava A, Sharma A, et al. Modified grey wolf optimization algorithm for transmission network expansion planning problem. Arabian Journal for Science and Engineering. 2018;43:2899. DOI: 10.1007/s13369-017-2967-3\n'},{id:"B26",body:'Glover F, Kochenberger GA. Handbook of Metaheuristics. Kluwer Academic Publishers; 2003\n'},{id:"B27",body:'Yang XS. Review of meta-heuristics and generalized evolutionary walk algorithm. International Journal of Bio-Inspired Computation. 2011;3(2):77-84\n'},{id:"B28",body:'Li Y, Gong G, Li N. Recent advances in modelling and optimizing complex systems based on intelligent algorithms. International Journal of Industrial Engineering: Theory, Applications and Practice. 2018;25(6):779-799\n'},{id:"B29",body:'Mladenovic N, Hansen P. Variable neighborhood search. Computers and Operations Research. 1997;24(11):1097-1100\n'},{id:"B30",body:'Hansen P, Mladenovic N. Variable neighborhood search: Principles and applications. European Journal of Operational Research. 2001;130:449-467\n'},{id:"B31",body:'Taglialenha SLS. Novas Aplicações de Meta heurísticas na Solução do Problema de Planejamento da Expansão do Sistema de Transmissão de Energia Elétrica [thesis]. 2008. DEE-FEIS-UNESP, Ilha Solteira\n'},{id:"B32",body:'Taglialenha SLS, Fernandes CWN, Silva VMD. Variable neighborhood search for transmission network expansion planning problem. In: Borsato M et al, editors. Transdisciplinary Engineering: Crossing Boundaries. ISPE TE. 2016;2016:543-552. DOI: 10.3233/978-1-61499-703-0-543\n'},{id:"B33",body:'Hansen P, Mladenovic N. A tutorial on variable neighbourhood search. Les Cahiers du GERAD, G-2003-46; 2003\n'},{id:"B34",body:'Haffner S, Monticelli A, Garcia A, Mantovani J, Romero R. Branch and bound algorithm for transmission system expansion planning using a transportation model. IEE Proceedings-Generation, Transmission and Distribution. 2000;147(3):149-156\n'},{id:"B35",body:'Oliveira GC, Costa APC, Binato S. Large scale transmission network planning using optimization and heuristic techniques. IEEE Transactions on Power Systems. 1995;10:1828-1834\n'},{id:"B36",body:'Risheng F, Hill DJ. A new strategy for transmission expansion in competitive electricity markets. IEEE Transactions on Power Systems. 2003;18(1):374-380\n'}],footnotes:[],contributors:[{corresp:"yes",contributorFullName:"Silvia Lopes de Sena Taglialenha",address:"s.taglialenha@ufsc.br",affiliation:'
Federal University of Santa Catarina, Brazil
'},{corresp:null,contributorFullName:"Rubén Augusto Romero Lázaro",address:null,affiliation:'
Electrical Engineering at FEIS-UNESP-Ilha Solteira, Solteira, Brazil
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