Comparison of the reference and proposed antennas .
In recent years, artificial meta‐surfaces, with the advantages of smaller physical space and less losses compared with three‐dimensional (3D) metamaterials (MTM), have intrigued a great impetus and been applied widely to cloaks, subwavelength planar lenses, holograms, etc. Typically, one most important part for meta‐surfaces’ applications is to improve the performance of antennas. In this chapter, we discuss our effort in exploring novel mechanisms of enhancing the antenna bandwidth using the magneto‐electro‐dielectric waveguided meta‐surface (MED‐WG‐MS), achieving circular polarization radiation through fractal meta‐surface, and also realizing beam manipulation using cascaded resonator layers, which is demonstrated from aspects of theoretical analysis, numerical calculation, and experimental measurement. The numerical and measured results coincide well with each other. Note that all designed antenna and microwave devices based on compact meta‐surfaces show advantages compared with the conventional cases.
- magneto‐electro‐dielectric waveguided meta‐surface
- fractal meta‐surface
- gradient phase meta‐surface
- beam manipulation
- bandwidth enhancement
Microstrip antennas have been used widely in recent wireless communication systems due to their coplanar structures, easy fabrication, and stable performances. However, conventional microstrip antennas face huge challenges, such as large size, narrow bandwidth, and inconvenient tuning of the working frequency.
Recently, artificial meta‐surfaces, planar inhomogeneous metamaterials composed of carefully selected elements with specific electromagnetic (EM) responses, have attracted much attention due to their strong abilities to control the wavefront of transmitted and reflected EM waves. With the unique EM properties, meta‐surfaces have found a lot of applications in focusing lens, cloaks, absorbers, antennas, and other microwave devices. One of the most important applications is to improve the performances of antenna, such as extending the working bandwidth and realizing circularly polarization radiation.
In Section 1, a new concept of planar magneto‐electro‐dielectric waveguided meta‐surface (MED‐WG‐MS) is introduced for the first time, which is capable of manipulating the effective permeability
2. Compact microstrip antenna with enhanced bandwidth using planar MED‐WG‐MS
Microstrip patch antennas are experimentally demonstrated with a lot of advantages, such as coplanar configuration, simple design, and low cost, which have widespread applications in a wireless communication system recently. However, the conventional microstrip antenna is electrically large, resulting from that the working frequency is exclusively dependent on the antenna size, which limited a further application of the patch antenna. Moreover, the antenna suffers from an intrinsically narrow bandwidth (BW). To address these issues, electromagnetic (EM) meta‐surfaces (MSs) have been proposed in recent years. Due to their strong EM abilities in improving the performances of conventional devices, MS antenna has become a research hotspot with remarkable achievements. However, using the MSs to increase the impedance BW of an electrically smaller microstrip patch antenna is still rarely reported.
Artificial magneto‐dielectric substrate has been verified to be a promising avenue to reduce the antenna size and extend the working band width [1–4]. A reported patch antenna, using a magneto‐dielectric substrate, achieves a relative 3.2% BW ordered by 6 dB return loss. However, the antenna suffers from a high profile, which is difficult for integration in compact devices. A magneto‐dielectric substrate using an embedded meander line (EML) is proposed in reference . A patch antenna based on this substrate realizes an antenna miniaturization and a BW improvement. Unfortunately, it needs an additional shield metal plate that complexes the fabrication. Therefore, it is an essential issue to design an MS antenna realizing simultaneously size reduction, BW extension, and also flexible frequency turning. In this section, we explored an improved strategy to simultaneously address aforementioned issues. An electrically smaller MS element is proposed by combining the electro‐dielectric and magneto‐dielectric waveguided substrates, defined as the magneto‐electro‐dielectric waveguided MS (MED‐WG‐MS) . The MED‐WG‐MS provides a freedom to control the wave impedance and refractive index simultaneously, achieving antenna miniaturization, bandwidth enhancement, and also flexible frequency modulation.
2.1. The concept and working mechanism of MED‐WG‐MS
As discussed in reference , planar WG‐MS is a special kind of artificial material residing in the planar waveguide environment, which is made up of an upper metallic layer and a lower ground plane. By introducing electric and magnetic resonators in the upper and lower metallic layers, MED‐WG‐MS is composed, which would bring about a series of interesting characteristics, including EM manipulation, further miniaturization, and also bandwidth enhancement. In addition, the MED‐WG‐MS is able to tune the working frequency easily, since more freedom is provided by the element. In other words, the antenna can work at any frequency just by manipulating the electric or magnetic resonators.
Then, we will discuss about the working mechanisms of the MED‐WG‐MS. There is a link between the patch size and the refractive index of the used substrate for a conventional patch antenna, which can be calculated as Eq. (1a) , where
Therefore, normalized with the conventional patch antenna, the compact factor (CF), defined as the ratio between the patch size for the conventional antenna and the MS antenna, can be derived from Eq. (1a) and (1b) as
The BW improving factor (BIF), evaluated by the BW ratio between the conventional antenna and the MS antenna, can be calculated as
Based on Eqs. (1)–(4) , CF and BIF can be controlled by manipulating the material parameters
2.2. The circuit model and EM property of MED‐WG‐MS
Based on the analysis in Section 1.1, we proposed a basic element as shown in Figure 1(a) . The element is a well‐known sandwich structure, consisting of an upper metallic two‐turn complementary spiral ring resonator (CSR) and a lower embedded Hilbert‐line (EHL) resonator, separated by a 1.5‐mm‐thick F4B spacer (dielectric constant
The meta‐surface, composing of a periodic element, is shined with a plane wave as the E‐field polarized along the
Next, we extracted the effective constitutive material parameters of four types of WG‐MSs using the standard retrieval process [10, 11] based on the calculated
2.3. Design of compact microstrip antenna with enhanced bandwidth
With a desirable MED‐WG‐MS element in hand, we can obtain effective parameters
Frequency tuning is essential in determining the antenna performances. In the following part, we will derive a three‐step frequency tuning method to achieve a flexible frequency tuning property. First, a coarse control over the operating frequency is considered by changing the outer radius
The bianisotropic response of the CSR has a large effect on the antenna behavior. By rotating the gap orientation
2.4. Numerical and experimental results
With the derived three‐step frequency tuning method, we can design an antenna operating exactly at frequency 3.5 GHz as shown in Figure 4. For experimental demonstration, we fabricate the antenna with the photograph shown in Figure 7 . The radiator of the antenna occupies an area of
Then, we evaluate the reflection coefficients of the proposed antenna and the reference antenna through the ME7808A vector network analyzer. Figure 6(c) illustrates the results of the reflection coefficients. There is a good agreement between the simulation and measurement for the designed antenna. Both the proposed antenna and the reference antenna operate exactly at 3.5 GHz, with the numerical (experimental) resonant dips as -25(-40) dB for the designed antenna and -21.5 dB for the reference one, respectively. The simulated (measured) 10 dB impedance bandwidth is about 115 (132) MHz for the designed antenna, corresponding to 3.29% (3.77%). However, it is 43 MHz for the reference antenna. A simply calculation indicates that the BW has been enhanced significantly by 207% for the designed antenna. Table 1 shows a detailed comparison between the two antennas, including the patch size, BW, and the values of CF and BIF. Note that the theoretical and simulated CF and BIF are in reasonable agreement with each other. The small CF = 0.57 and large BIF = 2.67 indicate that the designed antenna has the best performance compared with previous magneto‐dielectric antennas [1–5]. The higher measured BIF is mainly ascribed to the random errors in measurement. To summarize, the proposed MED‐WG‐MS element, coupled with the derived three‐step working frequency method provide a guideline to design the antenna with a bandwidth enhancement at any frequency. In addition, the fabrication of the designed antenna is more convenient without using parasitic elements or metallic via holes [13, 14].
|Type||Size(mm2)||BW(MHz (%))||CF||BIF||Antenna efficiency|
|Conventional patch antenna||25.4 × 27.4||43 (1.23%)||0.41/0.57||2.87/2.67 (3.07)||95.68%|
|Our work ||20 × 20||115/132 (3.29/3.77%)||93.23%|
There is a link between the slope of the input admittance and the BW for an antenna. A flatter response of the input admittance means a wide impedance‐matching property. Figure 8 shows comparison of the input admittance for the designed antenna and the reference antenna . It is worth noting that both the real part and the imaginary part of the input admittance for the designed antenna show a flatter response than that of the reference antenna, indicating an enhanced BW for the designed antenna.
Next, we evaluate the field distributions for both the designed antenna and the reference one. Figure 9(a) and (b) shows the comparison of the H‐field distributions for both antennas . There is a nearly periodic distribution for the H‐field of the proposed antenna, which is attributed to the loading of the MED‐WG‐MS cells. More importantly, based on the material parameters, there is larger wave impedance √ _______ μ eff / ε eff for the designed antenna. The larger wave impedance induces a weaker H‐field intensity and a smaller quality factor, which explained the enhancement of the BW from another aspect. The field intensity along the line
Finally, we examine the far‐field radiation performances by HFSS simulation and measurement in an anechoic chamber. The three‐dimensional (3D) radiation patterns at three representative frequencies (the lower frequency 3.440 GHz, the center frequency 3.5 GHz, and upper frequency 3.555 GHz) are depicted in Figure 10(a)–(c) . Given the defects in the ground plane, seemingly bidirectional radiation patterns are observed for the three frequencies. But consistent with the conventional patch antenna, the designed one still works at TM10 mode. The radiation patterns in both principal planes at 3.5 GHz are measured, as the results shown in Figure 10(d). The level of the cross‐polarization is better than -22.8 dB. Referring to the antenna gain in Figure 10(e), good agreement is observed for the simulated and measured results. Within the 10 dB impedance bandwidth, the gain is higher than 4.6 dBi. Moreover, the radiation efficiency of the antenna is about 93.23%.
In summary, a new strategy of enhancing the BW and reducing the size of a patch antenna is proposed by simultaneously manipulating the material parameters
3. Miniaturized circularly polarized antenna with fractal meta‐surface and fractal resonator
We have discussed about the strategy of employing MED‐WG‐MS element to manipulate the effective material parameters
Unlike linearly polarized (LP) antennas, microstrip circularly polarized (CP) antennas have a stable date transmission rate regardless of the polarizations of the transmitter and the receiver, thus have found numerous applications in the recent wireless communication system [15, 16]. However, traditional CP antennas are designed by involving various perturbations (such as truncated corners), which conflicts with the miniaturization requirement and wide bandwidth applications [17, 18]. In recent years, meta‐surfaces, such as reactive impedance surface (RIS), have been applied to enhance the performances of CP antennas [19–28], such as realizing miniaturization by loading RIS , achieving multi‐frequency operation [21, 25]. However, there are never open reported techniques to reduce the profile of the CP antenna. In this section, fractal concept has been introduced to design RIS, which realizes both miniaturization and low profile. Good performances of the designed CP antenna by using the fractal RIS are numerically and experimentally demonstrated.
First, we show the perspective view of the proposed Hilbert fractal RIS (HRIS)‐inspired Wunderlich‐shaped fractal complementary split ring resonator (WCSRR)‐loaded CP antenna , as shown in Figure 11 . The antenna consists of three layers, the upper metallic radiator, the HRIS spacer, and the lower metallic ground plane. WCSRR is etched in the metallic radiator to realize both antenna miniaturization and CP wave. 6 × 6 HRIS elements are loaded on the F4B substrate under the patch radiator. Two inexpensive F4B dielectric layers (
3.1. The EM property of the fractal RIS
RIS, proposed by Kamal et al. , has been used widely in antennas to reduce the radiator size and enhance the working bandwidth. In Figure 12(a) , conventional RIS (CRIS) consists of a periodic system of metallic patches (
To evaluate the EM response of the HRIS, Figure 14(a) depicts the reflection phases of HRIS with different IOs. Note that the fair comparison is made with the geometrical parameters consistent except the IOs of the HRISs. At the resonant frequency, the reflection phase is about zero . As seen from Figure 14(a), the operating frequency changes from 9.1 to 7.41 GHz as IO increases from 0 to 2. The second HRIS cell occupies a dimension of λ0/8.1 × λ0/9 × λ0/16.2, which is much smaller than the working wavelength. With a larger IO of the HRIS, stronger space‐filling property of the structure is obtained, which induces a smaller resonant frequency. Here, IO = 2 is selected from the tradeoff of an easy fabrication and an electrically smaller structures. Figure 14(b) illustrates the dependence of reflection phase for RISs on the substrate thickness
3.2. Working principle of compact fractal resonator
With a unique EM property and a planar structure, split ring resonator (SRR), and its complementary part, CSRR have been utilized in designing microwave devices and enhancing their performances. However, it is still a great challenge to excite CP wave by integrating the fractal strategy with the CSRR structure. We combine the Wunderlich‐shaped fractal structure and the CSRR (WCSRR) to achieve not only antenna miniaturization but also good CP radiation. To demonstrate the advantage of WCSRR over the traditional CSRR, the EM response of three kinds of CSRRs is compared. Figure 15 presents the basic topologies of different SRRs, such as conventional SRR, meander‐line‐loaded SRR (MSRR), and WSRR. Based on the Babinet principle, we etched the CSRR, MCSRR, and WCSRR structures in the patch radiator. The reflection coefficients and axial ratio (AR) for antennas with different slots are provided in Figure 16. Good impedance matching property for all cases can be observed clearly by the resonant dips, as shown in Figure 16(a). The conventional antenna without CSRR loading operates at about 3.8 GHz, whereas the center frequency reduces significantly to 3.75, 3.65, and 3.5 GHz for the CSRR‐, MCSRR‐, and WCSRR‐loaded antennas, respectively. The extending of the current path caused by the fractal curve induces the considerable reduction of operating frequency, which is equivalent to realize antenna miniaturization working at the same frequency. Figure 16(b) shows the effects of CSRRs on the AR performances. Note that the slots have a significant effect on the antenna AR. An obvious linearly polarized antenna is obtained without loading CSRRs. The values of AR reduce significantly by loading CSRRs. The best performance of the CP antenna is observed by loading WCSRR, which achieves a lower operating frequency and a smaller AR value. Therefore, we can use the Wunderlich‐shaped fractal slot to improve CP antenna performances.
3.3. Numerical results of a CP antenna
The feeding position as well as the slot position is very important in determining both input impedance and AR value. Antenna performances are simulated by HFSS via tuning the feeding position. The parameter
Then, we study the electric field distribution of the CP wave at 3.5 GHz, as the results shown in Figure 18. The electric field is mainly concentrates on the radiation patch and the WCSRR slot. Left‐handed CP wave is observed clearly with a clockwise‐rotated electric field. The variation of the field on the WCSRR indicates that the slot plays an essential role in exciting the CP wave. Right‐handed CP wave can be excited by changing the position of the slot.
3.4. Fabrication and experimental results
For experimental demonstration, the finally double‐layered CP antenna is fabricated based on a standard printed‐circuit‐board (PCB) technology, with the fabricated sample for the upper layer and the lower layer shown in Figure 19. The two layers with the same footprints are boned tightly together by an adhesive. Then we experimentally evaluate the performances of the CP antenna.
First, we examine the impedance matching property of the antenna. The reflection coefficient of the sample is measured by a vector network analyzer (ME7808A). Figure 20(a) depicts a comparison of the simulated and measured reflection coefficients. We can see clearly that there is an excellent agreement between the numerical and experimental results. A slight frequency shift upward about 20 MHz in the experiment is mainly attributed to the fabrication errors which is inherent and the introduction of the adhesive in the assembly process. The simulated (measured) reflection coefficient has a resonant dip of -22 dB (-18 dB) at about 3.5 GHz (3.52 GHz), respectively. The working bandwidth, characterized by the 10 dB return loss, is about 132 and 127 MHz for the simulation and measurement, respectively. The relative BW is calculated of 3.77% and 3.61%.
Then, we evaluate the AR ratio of the designed CP antenna. In the experiment, the AR is measured by the intensity ratio between
Finally, we measure the far‐field radiation patterns in
In summary, we have proposed a new strategy to design CP antenna by combining a fractal RIS and a fractal slot. For proof of the strategy, a CP antenna by loading HRIS structure and the WCSRR slot is designed, assembled, and measured. The experimental result coincides well with the simulated case. The antenna advances in many aspects such as a compact size (40 mm × 45 mm × 2.5 mm), good AR property (0.75 dB), and comparable radiation gain (6.3 dBic). In addition, the antenna is fabricated based on the PCB technology, free of via holes and without using the complex feeding network.
4. Ultra‐thin polarization beam splitter using TGMS
Meta‐surfaces, as a 2D planar inhomogeneous metamaterials, are composed of carefully selected elements with specific EM responses, have attracted much attention recently due to their strong abilities to manipulate the wavefront of transmitted and reflected EM waves. Very recently, scientists and engineers have designed functional devices by using meta‐surfaces. One of the most important applications is to realize the polarization beam splitter (PBS). PBS is a typical device to manipulate the differently polarized waves independently, which has been found essential in photonics. Conventional PBSs, achieving by natural crystal birefringence  and 2D photonic crystals [31, 32], suffer from low efficiencies and limited splitter angle. The PBS performances have been enhanced by using the semiconductor meta‐surface , photonic‐integrated circuits , and 2D metamaterials [35–37]. However, these devices are electrically large and complex in fabrication. In this subsection, we propose a novel PBS based on the 2D transmissive phase gradient meta‐surface (TPGM) . The proposed PBS is designed based on the generalized Snell's laws, which has a low‐profile about 0.1 λ0, high transmission efficiency, and also convenient wave control.
4.1. Polarization‐controlled mechanism of local element
According to the geometrical optics, the EM wave would be reflected or refracted when passing through the interface of two media. When the EM wave propagates in a uniform medium, the wave vector can be written as
The generalized laws of reflection and refraction, derived from the Fermat's principle, indicate that the anomalous reflection or refraction phenomenon can be observed at the interface when the phase discontinuity exists . At the same time, the EM response of the GMS is independent when excited with differently polarized waves. Illuminated by a normally incident EM wave along the
where the superscript
For the practical GMS, the phase gradient can be calculated as
According to Eq. (7), two points should be highlighted. First, the phase distribution
Based on Eq. (7), a transmissive PBS can be realized by controlling the parameters ξ x(x ) and ξ y(y ) independently. In other words, we can manipulate the transmission phase
To make the principle of the PBS clear, Figure 22 provides the schematics of anomalous refractive phenomena in four different conditions. 2D TPGMs, positioned at the
4.2. High‐efficiency transparent principle of the transmissive element
For the first step, we should find a transparent element not only with a high transmission coefficient but also a changeable transmission phase. In general, a complete phase shift range over 360° is required to guarantee a free wavefront control. Note that EM coupling among the cascaded layers can enlarge the transmission variation range and improve the transmission coefficient [40–44]. Here, three‐layer cascaded element is chosen as the basic element, as the topology shown in Figure 23(a). The basic element consists of three identical metallic layers and two intermediate dielectric layers. The commonly F4B substrate is used with a thickness
To obtain the transmission spectra of the basic element, we chose a typical element with
4.3. Design and analysis of 2D TGMS
With a well‐designed element in hand, we can design the TPGM by carefully chosen the elements satisfying the phases both for horizontal and vertical polarizations. A systematical study on the TPGM can provide a guideline for the upcoming PBS design. We derive a four‐step design procedure for the general TPGM. For the first step, a transmissive element with a high transmission coefficient and changeable transmission phase should be obtained. Second, the phase distributions at the position (
Based on derived design procedure, four kinds of TPGM are designed based on the phase distributions, as shown in Figure 22. Figure 26 provides the electric fields distributions of E// and
4.4. Fabrication and evaluation of PBS
With a well‐designed TPGM4, we can further optimize its performance and build a novel PBS by launching the meta‐surface by a horn antenna working at the X band. Figure 27 shows the photograph of the fabricated sample. We can see that 24 × 24 unit cells are adopted for the final PBS, which occupies a total volume of 256 × 256 × 3 mm3, corresponding to 6.6 λ0 × 6.6 λ0 × 0.1 λ0. It is worth noting that the thickness of the PBS is much smaller than the reported cases. Moreover, the PBS has no complex structures and is fabricated based on the simple print circuit board (PCB) technology. The PBS consists of 4 × 4 super unit cells, as the top and side views of the super unit cell are shown in Figure 27(a) and (b). Compared with the working mechanism of the previous PBSs, the designed one is intuitionistic and simple. We can control the differently polarized waves independently due to the different phase gradients at
The impedance matching property of the PBS plays an essential role in determining the working efficiency. We evaluate the reflection coefficients of the PBS by launching with different polarizations. First, we assemble the fabricated sample. We used a 20‐mm‐thick foam plate to support the TPGM and ensure the length
Then, we examine the 3D far‐field patterns of the novel PBS at its working frequency of 10 GHz under excitation of differently polarized waves. Without the TGMS, the horn antenna radiates a narrow beam along the
Next, we evaluate the 2D radiation patterns of the novel PBS through the far‐field measurement system in an anechoic chamber. Here, three excitation polarizations are considered to investigate the far field performances of the designed PBS, as the simulated and measured results shown in Figure 30. The measured results coincide well with the simulation except the radiation pattern in the
The polarization splitting ratio, defined as the radiation gain between two separated radiation beams, is a very important factor for the PBS. We simulate and measure the radiation patterns at the conical surface
In summary, we propose a new strategy to design the PBS aiming at reducing the structure thickness, realizing good polarization splitting ratio, and also high efficiency. A TPGM is proposed based on a carefully designed element, which realizes a high transmission coefficient of more than 0.8, a wide phase variation range over 330° and also a polarization‐independent property. The TPGM achieves good beam separation performances as excited by different polarizations. Launched by a wideband horn antenna with a suitable length, an ultra‐thin PBS is designed, fabricated, assembled, and measured. Numerical and experimental results show that the PBS can deflect incident waves with different polarizations to different directions with high polarization splitting ratio.
In this chapter, we have reviewed our recent efforts in utilizing electrically small meta‐surface elements to improve antenna performances and design functional devices, among which three most important aspects have been investigated in depth. First, the constitutive material parameters are controlled by the proposed MED‐WG‐MS elements, achieving a compact microstrip antenna with enhanced bandwidth. Second, fractal meta‐surface and fractal resonator are combined to achieve a CP antenna with a low profile. Third, a high‐performance PBS has been proposed based on the TGMS, which shows advances in many aspects such as separating and controlling the orthogonally polarized waves with a polarized splitting ratio better than 18 dB, obtaining a comparable bandwidth of more than 600 MHz, and also gaining high transmission efficiency. Our results pave a new avenue for both engineers and scientists to realize their devices or demonstrate their findings.
This work was supported by the National Natural Science Foundation China under Grant Nos. 61372034, and 61501499 and also the Natural Science Foundation of Shaanxi Province under Grant Nos. 2016JM6063 and 2016JQ6001.
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