The optimized antennas (all dimensions in mm) [27].
\r\n\tLiterature showed the presence of ACE2 receptors on the membrane of erythrocyte or red blood cell (RBC), indicating that erythrocyte (RBC) can be considered as a peripheral biomarker for SARS-C0V2 infection.
\r\n\r\n\tIncreased levels of glycolysis and fragmentation of RBC membrane proteins were observed in the SARS-C0V2 infected patients, demonstrating that not only RBC’s metabolism and proteome but its membrane lipidome could be influenced by SARS-C0V2 infection changing the homeostasis of the infected erythrocyte. This altered RBC may result in the clot and thrombus formation; the major signs of critically ill Covid-19 patients.
\r\n\r\n\tThis book is going to be a succinct source of knowledge not only for the specialists, researchers, academics and the students in this area but for the general public who are concern about the present situation and are interested in knowing about simple non-invasive measures for identifying viral and bacterial infections through their red blood cells.
",isbn:"978-1-83969-121-8",printIsbn:"978-1-83969-120-1",pdfIsbn:"978-1-83969-122-5",doi:null,price:0,priceEur:0,priceUsd:0,slug:null,numberOfPages:0,isOpenForSubmission:!0,hash:"fa5f4b6ef59e28b6e7c1a739c57c5d2f",bookSignature:"Prof. Kaneez Fatima Shad",publishedDate:null,coverURL:"https://cdn.intechopen.com/books/images_new/10494.jpg",keywords:"Spike Protein, Hemoglobin, Proteins for Oxygen Transport, Altered Protein Structures, RBC ACE Receptors, RBC ACE-2 Receptors, Carboxypeptidase, Mas Receptor, Metabolomics, Gas Transport, Glucose-6-Phosphate, Phosphoglycerate",numberOfDownloads:null,numberOfWosCitations:0,numberOfCrossrefCitations:null,numberOfDimensionsCitations:null,numberOfTotalCitations:null,isAvailableForWebshopOrdering:!0,dateEndFirstStepPublish:"October 15th 2020",dateEndSecondStepPublish:"November 30th 2020",dateEndThirdStepPublish:"January 29th 2021",dateEndFourthStepPublish:"April 19th 2021",dateEndFifthStepPublish:"June 18th 2021",remainingDaysToSecondStep:"2 months",secondStepPassed:!0,currentStepOfPublishingProcess:3,editedByType:null,kuFlag:!1,biosketch:"Dr. Shad is a governing body member and mentor of Women in World Neuroscience (WWN), a division of the International Brain Research Organization (IBRO). She is also a member of IBRO-APRC Global Advocacy responsible for brain research funding distribution in this region.",coeditorOneBiosketch:null,coeditorTwoBiosketch:null,coeditorThreeBiosketch:null,coeditorFourBiosketch:null,coeditorFiveBiosketch:null,editors:[{id:"31988",title:"Prof.",name:"Kaneez",middleName:null,surname:"Fatima Shad",slug:"kaneez-fatima-shad",fullName:"Kaneez Fatima Shad",profilePictureURL:"https://mts.intechopen.com/storage/users/31988/images/system/31988.jpg",biography:"Professor Kaneez Fatima Shad, a neuroscientist with a medical background, received Ph.D. in 1994 from the Faculty of Medicine, UNSW, Australia, followed by a post-doc at the Allegheny University of Health Sciences, Philadelphia, USA. She taught Medical and Biological Sciences in various universities in Australia, the USA, UAE, Bahrain, Pakistan, and Brunei. During this period, she was also engaged in doing research by getting local and international grants (total of over 3.3 million USD) and translating them into products such as a rapid diagnostic test for stroke and other vascular disorders. She published over 60 articles in refereed journals, edited 8 books, and wrote 7 book chapters, presented at 97 international conferences, mentored 34 postgraduate students. Set up a company Shad Diagnostics for the development of cerebrovascular handheld diagnostic tool Stroke meter into a wearable.",institutionString:"University of Technology Sydney",position:null,outsideEditionCount:0,totalCites:0,totalAuthoredChapters:"4",totalChapterViews:"0",totalEditedBooks:"6",institution:{name:"University of Technology Sydney",institutionURL:null,country:{name:"Australia"}}}],coeditorOne:null,coeditorTwo:null,coeditorThree:null,coeditorFour:null,coeditorFive:null,topics:[{id:"16",title:"Medicine",slug:"medicine"}],chapters:null,productType:{id:"1",title:"Edited Volume",chapterContentType:"chapter",authoredCaption:"Edited by"},personalPublishingAssistant:{id:"280415",firstName:"Josip",lastName:"Knapic",middleName:null,title:"Mr.",imageUrl:"https://mts.intechopen.com/storage/users/280415/images/8050_n.jpg",email:"josip@intechopen.com",biography:"As an Author Service Manager my responsibilities include monitoring and facilitating all publishing activities for authors and editors. From chapter submission and review, to approval and revision, copy-editing and design, until final publication, I work closely with authors and editors to ensure a simple and easy publishing process. I maintain constant and effective communication with authors, editors and reviewers, which allows for a level of personal support that enables contributors to fully commit and concentrate on the chapters they are writing, editing, or reviewing. I assist authors in the preparation of their full chapter submissions and track important deadlines and ensure they are met. I help to coordinate internal processes such as linguistic review, and monitor the technical aspects of the process. As an ASM I am also involved in the acquisition of editors. 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Knobology is a terminology that describes the manipulation of ultrasound knobs and system controls in order to obtain the best image possible from diagnostic ultrasound. The inadequate use of knobology variables may impair image quality and can result in misdiagnosis.
\nThis chapter explains the functions of the various ultrasound system controls and knobs and the impact they have on greyscale ultrasound imaging. It demonstrates the effect of transducer selection on image quality, and the role of knobology variables in image optimization. This includes a description of the Application Preset, Output Power, Overall Gain, Time Gain Compensation (TGC), Focus, Depth, Zoom, Dynamic Range and Tissue Harmonics. The influence of these essential knobs and system controls on spatial resolution (including lateral and axial resolution) and Contrast resolution are explained. In addition, the utility of Doppler knobs for imaging abdominal blood vessels are also explained and demonstrated.
\nThe need to adhere to the principle of As Low As Reasonably Achievable (ALARA) is also explained with emphasis on the imaging of neonates and children. Lastly, the chapter also emphasizes the potential detrimental effect of underutilizing ultrasound knobs and system controls in abdominal sonography.
\nSwitching on the ultrasound machine is the first knob to press if the machine is switched off. By switching on the machine, the ultrasound system is given access to a source of electricity, which excites the tiny piezoelectric crystals within the connected transducer. These piezoelectric crystals emit sound waves as a result of their exposure to electricity. The sound waves produced by the piezoelectric crystals can then be transmitted into the human body, normally aided by a coupling gel which serves as an acoustic medium for eliminating the air between the surface of the transducer and the skin.
\nModern machines allow the operator to preset an application setting for a certain examination type. Ultrasound imaging is used for a wide range of medical applications. Aside its use in assessing the abdomen, it is also used in obstetrics and gynecology, cardiac and vascular examinations, and other small-part examinations such as breast, thyroid, and musculoskeletal imaging. Different sonographic settings are needed for the various examinations, due to their differences in terms of the depth of region of interest, tissue-type, and the size of organs and structures in that region. Because of the uniqueness of these examinations, adjusting the settings between patients for a different examination can be time consuming, and may compromise the adherence to ALARA principles. In addressing this limitation, the manufacturer makes it easier by allowing the operator to select the type of examination which will activate the pre-defined factory settings for the specific type of examination. By selecting the appropriate ‘Application Preset’ for abdominal examination, the pre-defined factory settings are activated for abdominal sonography. This automatically adjusts the basic settings for the selected examination, which include an adjustment of the transducer frequency, acoustic Output Power, Overall Gain, Dynamic Range, Depth and other related settings.
\nPerforming an abdominal ultrasound with a different application preset may impair the image quality which could mislead image interpretation. For example, a user performing an obstetric examination may identify a need for including abdominal examination without switching to the abdomen preset. This may impair the image quality of the abdominal examination if careful adjustments of relevant knobology variables are not made. In \nFigure 1a\n, obstetric preset was used in imaging the kidneys of an obstetric patient who complained of flank pain during an obstetric ultrasound examination. Upon using the basic obstetric preset without further manipulation of essential knobs, there was the tendency of suspecting a focal lesion in the right kidney (see arrow in \nFigure 1a\n). However, a switch to the basic abdomen preset without further manipulation resulted in an improved image quality which shows a normal kidney (see arrow of \nFigure 1b\n in the same person). It therefore suggests that much more manipulation of knobs will be required for selecting the ‘wrong’ application preset which may unduly extend the duration of the examination as a compromise on ALARA principles.
\n(a) Image of the right kidney with OB preset suggests a focal change within kidney (see arrow). (b) Image of the right kidney with abdomen preset suggests normal appearance (see arrow).
Ultrasound images are produced from high frequency sound waves that are emitted by the transducer, typically in the range of 1–15 MHz [1, 2]. The frequency of the transducer is determined by the thickness of the piezoelectric crystals and the damping material behind them [3]. In producing a higher frequency, the manufacturer places a damping material behind very thin piezoelectric crystals in order to shorten the pulses of sound waves that are emitted [3]. However, shorter pulses of sound waves are unable to penetrate deeper because of shorter wavelength [3]. Due to this penetration limitation, different types of transducers are designed with different ranges of frequency. Higher frequency transducers offer better resolution at the expense of depth penetration, whilst lower frequency transducers offer better depth penetration for poorer image resolution [2, 3].
\nSince most abdominal organs such as the liver, spleen, kidneys, pancreas and aorta are relatively deeper, lower frequency transducers are used for this type of examination. Unlike the transducers designed for other examinations, the transducers for abdominal examination (i.e. sector or curvilinear) have a divergent and wider far field. Aside the lower frequency of curvilinear and sector transducers which makes image resolution relatively poorer, there is also an increase in attenuation as the sound beam travels deeper. This may adversely affect the image resolution of abdominal sonography. It is therefore incumbent on the operator to make a careful choice between better image resolution and depth penetration.
\nThe typical frequency range for curvilinear transducers is in the range of 2-5 MHz. In selecting a frequency for an abdominal examination, the operator should consider the size of the patient. If the patient is smaller in size, a higher frequency should be used for better spatial resolution. Particularly in neonates and children, a higher frequency is highly useful, as this is likely to produce better image resolution to shorten the duration of the examination in fulfillment of ALARA principles. Secondly, children are less likely to cooperate during the examination, therefore using a lower frequency such as 3 MHz for abdominal examination may unduly delay the examination because of the lack of patient cooperation and a poorer image resolution. \nFigure 2a\n and \nb\n demonstrates two images of the right and left kidneys obtained from a 3-year-old infant with the higher frequency obviously showing more details than the lower frequency. However, a low frequency of 3-4 MHz is often ideal for imaging the average-sized adult, whilst larger or much more obese adults may require as low as 2 MHz of frequency for adequate depth penetration.
\n(a) Image of the right kidney of right and left kidney in a 3-year-old non-cooperating patient showing poorer image resolution because of lower transducer frequency of 2.5 MHz. (b) Image of the right kidney of right and left kidney in the 3-year-old non-cooperating patient showing better visualization of renal margins because of lower transducer frequency of 2.5 MHz.
In addition, a linear transducer may also be used during abdominal ultrasound. Linear transducers use higher frequencies for imaging structures that are more superficial, such as the anterior abdominal wall and the surface of the liver. They are also used in assessing the appendix.
\nThe acoustic output power of the machine must be considered at all times by the operator. As indicated above, selecting the appropriate preset for abdomen ultrasound will automatically adjust the output power to the recommended level. However, while it is important to observe the ALARA principle by using the minimum output power possible, the operator must not compromise image quality for output power reduction which may lead to misdiagnosis. In essence, there should be a balance between maximizing image quality with the minimum output power possible as a measure for reducing the risk of biological effect. Usually, the ultrasound machine will display the output power on the screen at all times, allowing the operator to be constantly informed (\nFigure 3a\n and \nb\n). However, while increasing the power output may be useful, it may also be needless in many cases. The over-all Gain can play a better and safer role in image quality optimization than the output power. \nFigure 3a\n and \nb\n demonstrate that there is no significant difference between the appearance the abdominal aorta if the output power is reduced by 50% and the overall gain is about 30 decibels.
\n(a) The appearance of the abdominal aorta at a reduced output power by 50%. (b) The appearance of the abdominal aorta when the output power increased was increased to 100 showed no significant difference %.
The overall gain is the recommended option to consider in place of increasing the output power. With the overall gain, image quality can be improved by adjusting the brightness of the entire field of view without increasing the intensity of transmitted sound energy. It achieves this by amplifying the echo-signals returning from the body after transmitting the sound waves. The overall gain can be considered as the ‘microphone’ in ultrasound imaging. The technology is similar to using a microphone to amplify someone’s voice for the listener. Increasing or decreasing the overall gain may improve contrast resolution for adequate visualization of the image. However, just as a microphone can sometimes produce noise and become a nuisance, increasing the overall gain beyond a certain point will affect contrast and spatial resolution by making the image appear too bright. Nonetheless, it is a knob you cannot do without in image optimization. Most modern machines integrate the overall gain in the Bmode or 2D knob, but it is still a separate knob in some machines. Manipulate the overall gain by adjusting it ‘up and down’ and carefully observe the changes that occur as you control the knob. \nFigure 4a\n and \nb\n shows images of adequate versus high overall gain and the effect it has on assessing the surface of the liver.
\n(a) Adequate overall gain of 31 decibels with liver surface showing. (b) Too high overall gain of 60 decibels with liver surface missing.
While the overall gain would adjust the brightness of the entire field of view, it may not address attenuation occurring at specific depths. Some structures in the body are much more affected by attenuation than others and would therefore need additional compensation for the loss of sound energy. For example, an optimum visualization of the left lobe of the liver requires a depth specific gain adjustment that is different from the gain compensation needed for optimum visualization of the right lobe. Hence the Time Gain Compensation (also known as Depth Gain Compensation), is a set of depth-specific slide controls that can be used for echo-signal amplification at different depths (see \nFigure 5\n). It allows the adjustment of echo-signals in the near-field, mid-field and far field to improve axial resolution. The TGC creates uniformity in the brightness of the echoes when used in conjunction with the overall gain. The best approach is to center all the TGC settings before adjusting the overall gain. After adjusting the overall gain, the TGC can then be adjusted to compensate for attenuation at specific depth.
\nTGC slide in the yellow circle.
During scanning, the system allows the operator to improve lateral resolution in a region of interest by adjusting the focal zone. This is an additional measure to minimize the effect of attenuation. However, while other controls such as the overall gain and TGC are effective for improving axial resolution, adjusting the focal zone is much more effective for improving lateral resolution. Lateral resolution refers to the ability to identify structures lying side-by-side as separate structures, while axial resolution refers to the ability to identify a structure lying on another structure as separate structures.
\nThe focal zone normally appears at the lateral side of Bmode as a triangular-shaped structure or a dot. It can be moved up or down by the operator and should be placed at the region of interest or posterior to that region. If a single focal zone is set too superficially a poorer image resolution will be observed in the far field (\nFigure 6\n). However if the focal zone placed below or at the level of region of interest, the resolution improves in the entire field of view (\nFigure 6\n). To improve lateral resolution in a wider region, more than one focal zones may be selected by the operator. However, increasing the number of focal zones also decreases the frame rate which has the tendency of slowing down the image production time to the detriment of temporal resolution. Thus using more focal zones slows down the scanning time which may not support the principles of ALARA in terms of keeping to a reasonable scanning time.
\nPoorer resolution when focal zone is positioned in the near is compared to focal zone positioned at the level of interest.
The Depth is special a knob for adjusting the distance of the field of view. Structures within the field of view can be moved far or closer by adjusting the Depth. This is to ensure that the region of interest is closer enough for optimum visualization. It is also to avoid showing regions that are not relevant to the area of interest. \nFigure 7a\n is an example of a far depth image of the pancreas, with a wide irrelevant space showing behind the spine. This irrelevant space can be avoided by adjusting the Depth closer for adequate visualization (\nFigure 7b\n). The structure of interest should always take the center stage by occupying about two-thirds of the field view. In order to avoid missing a pathology beyond the field of view, the best practice is to adjust the Depth for a far field of view before adjusting for a closer field of view. \nFigure 8\n is an example of how one can miss a pathology, if the Depth is not adjusted for adequate visualization beyond the field view. It demonstrates how a closer Depth would have missed the pleural effusion if a far depth image was not assessed.
\n(a) Far depth. (b) Closer depth.
Missing information on the right-side because of depth adjustment.
However, while moving the depth closer and far is necessary for evaluating various conditions and ruling out pathologies, moving the Depth closer has the tendency of generating noise which can worsen contrast resolution and may even mimic a pathology.
\nThe zoom is used for magnifying the area of interest. Unlike the depth which magnifies by moving the area of interest closer, the zoom actually magnifies by making the region of interest appear bigger. Another limitation of the depth that is addressed by the zoom is the ability to enlarge a specific region of interest. Without using the zoom, measuring some tiny structures may difficult because of poor spatial resolution. For instance, in measuring the thickness of the gallbladder wall, using the zoom improves the visualization of the wall for an accurate measurement (\nFigure 9a\n and \nb\n).
\n(a) Read zoom. (b) Write zoom.
Some manufacturers use READ zoom for their magnification, while others use WRITE zoom. Both read zoom and write zoom can produce poorer image depending on the size of the area magnified. However, READ zoom produces the worse kind of images because it relies on stored images which enlarges the pixel density in that region (\nFigure 9a\n). On the other hand, WRITE zoom tries to maintain the pixel density by zooming the image live which produces a better spatial resolution. Operators should check the type of zoom in their machine in order to appreciate how much zooming can be done without compromising the image quality.
\nThe Dynamic Range is a control on the ultrasound system that allows the operator to determine the range of shades of gray to be displayed on the monitor. Broad shades of gray displays a wider range of echo-intensity between bright and dark and produces a smoother image overall, whilst narrow shades of gray displays a narrower range of echo-intensity between bright and dark and produces a higher contrast between two regions of different echogenicity. In abdominal sonography, a broad dynamic range is the most appropriate option for assessing the echotexture of homogeneous soft-tissue structures like the liver, pancreas and spleen. Narrow dynamic range is most appropriate for assessing anechoic structures such as the aorta and IVC. \nFigure 10a\n shows the effect of narrow dynamic range of the pancreas in comparison to the liver, and \nFigure 10b\n shows the effect of broad dynamic range on the pancreas which shows poor differentiation in echotexture in comparison to the liver. In \nFigure 11\n also shows the effect of long and short dynamic range on the appearance of the IVC.
\n(a) Narrow dynamic range. (b) Broad dynamic range.
Broad versus narrow dynamic range of IVC.
Tissue harmonic Imaging (THI) is an additional control for image optimization in most ultrasound machines. It improves image quality by eliminating weak echoes that cloud the image when the fundamental frequency of the transducer is used. It replaces the returning echoes from the fundamental frequency with echoes in the harmonic frequency which improves spatial resolution. This eliminates side lobe artifacts and noticeable noise in the area of interest. It can therefore be used in conjunction with the utilization of other knobs that may generate noise. For example, noise generated by increasing the Depth can be instantly eliminated by activating THI. \nFigure 12\n also shows an increase in noise as a result of increasing the overall gain and Depth, and how it is instantly eliminated by the activation of THI. The activation of the THI in \nFigure 12\n instantly changed the settings from the fundamental frequency to the harmonic frequency. In \nFigure 13\n, you also appreciate the importance of THI, in terms of how it improves visualization of the margins of liver surface in comparison with the adjacent image which did not use THI.
\nNoise from increased overall gain and depth is instantly eliminated THI activation.
Improved liver margins with activated THI.
The ultrasound machine also has a freeze button which enables the operator to stop and evaluate the image quality before storage. Saving an image without freezing implies that the image was not evaluated for quality. Freezing the image before storage is therefore recommended.
\nThe cineloop is additional control that helps with selecting the best of the image frozen image. It displays image frames acquired in the last few seconds prior to freezing. The cineloop can be highly useful when scanning children.
\nAn abdominal ultrasound examination may also require the assessment of blood vessels and Doppler evaluation of blood flow. The fundamental knobs that influence both color and spectral Doppler imaging include the Doppler gain, pulse repetition period (PRF), and the wall filter. In assessing the presence of flow in smaller blood vessel, the minimum standard is to adjust the system for a higher Doppler gain, a lower PRF and a lower wall filter [4]. Careful manipulation is used in balancing these knobs, as a slight overlap between them can generate noise artifacts.
\n\n\nFigure 14a\n shows the poorer flow in the hepatic vein in comparison to the portal vein as a result of higher PRF. This is much improved in \nFigure 14b\n with decreased PRF.
\n(a) High PRF with low flow sensitivity in hepatic vein. (b) Low PRF with high flow sensitivity in hepatic vein.
In larger abdominal blood vessels such as the aorta, additional knob controls that are highly relevant include the imaging angle which must not be parallel to the surface of the transducer. \nFigure 15a\n and \nb\n shows the effect of imaging angle on color flow in the aorta which is absent when the vessel is parallel to the surface of the transducer.
\n(a) Color flow showing in angled vessel. (b) Color flow absent in parallel vessel.
In spectral Doppler Imaging, however, lower PRF may cause aliasing artifact, especially when the baseline is high [5]. This can be corrected by increasing the PRF of the spectral waveform and lowering the baseline. \nFigure 16a\n shows aliasing artifact of the Superior Mesenteric Artery (SMA) which was as a result of a lower PRF and a higher baseline. By increasing the PRF and lowering the baseline, a normal waveform of the SMA was obtained in \nFigure 16b\n. Other essential knobology settings which improves spectral waveform in the assessment of peak systolic velocity include using a smaller sample gate and ensuring an angle correct setting that aligns with the vessel wall as demonstrated in \nFigure 16a\n and \nb\n.
\n(a) Aliasing artifact in the superior mesenteric artery as a result of lower PRF and higher baseline. (b) Adequate waveform for assessing peak systolic velocity in the superior mesenteric artery, after increasing the PRF and lowering the baseline.
Understanding the influence of knobology in ultrasound imaging is essential in abdominal sonography. The image quality can be optimized by selecting the appropriate application preset and transducer frequency. While using the highest output power may be useful, it is not necessary in many instances. The various knobology variables with direct influence on greyscale and color Doppler should be regularly manipulated by the operator for the best image possible in abdominal sonography.
\nIn recent years, UWB technology has mostly focused on consumer electronics and wireless communications. Federal Communication Commission (FCC) issued a report in February 2002, allowing the commercial and unlicensed deployment of UWB applications in USA for both indoor and outdoor spectral mask. This wide frequency allocation initiated a lot of researches from both industry and academia [1]. UWB is used for short and medium range of radio communications and positioning applications.
The European regulatory body issue similar restrictions are shown in Figure 1. The key limitations for wireless communication using UWB are mentioned in [2, 3, 4].
European regulatory body spectrum.
UWB impulse radio system has several advantages over other conventional systems [5].
High data rate wireless transmission: UWB systems can support more than 500 Mb/s data transmission rate in the range of 10 m, which enables for new services and applications.
High precision ranging: UWB systems have good time-domain resolution and it could be provided centimeter accuracy for location and tracking applications.
UWB is used for location and tracking applications with cm accuracy.
UWB can operate under LOS and NLOS environments for signal penetrating obstacles.
UWB system is capable of resistance to multipath fading.
The power spectral density is very low so it is secure and can coexist with other services such as WLAN, GPS, cellular system, etc.
The UWB system has low cost due to using CMOS technology.
Due to UWB technology and using nanosecond pulses in many applications such as military and biotechnology applications [6, 7, 8], the need for very broadband circularly polarized antenna has emerged. These UWB CP antennas are the substitution of the narrowband CP microstrip patch antennas [8, 9, 10].
One of the most commonly used devices to control the spectrum of radio frequency signals and necessary in an UWB radio system whether in impulse system or multiband system, in order to reconfigure the UWB signal to satisfy the spectrum regulation is the filter. Ultra-wide band (UWB) band-pass filter that achieves ultra bandwidth from 3.1 to 10.6, low insertion loss, low and flat group delay, out-band performance can be considered a well-designed UWB band-pass filter.
In UWB band-pass filters, one can use many techniques in their design such as composite low-pass and high-pass structure [11], multiple-mode resonator structure [14], and short-circuited stub [12, 13]. Because UBW components occupy a large bandwidth which may be extended from 3.1 up to 10.6 GHz, interference attenuation due to coexisting services should be avoided. This is the motivation of using switchable or tunable narrow band notch within the passband of the UWB filter [15, 16, 17]. This may be achieved by many methods such as using additional notch resonators [18], embedded open stubs [19], asymmetric coupled fed lines [20], out-of-phase transmission cancelation [21], meander-line slots [22], and short-circuited stub resonators in a multilayer periodical structure [23].
To deal with different co-existed communication needs, the reconfigurable notch band implementation is required, but little research is concentrated on a UWB BPF with reconfigurable, switchable, or tunable notch bands as in [24, 25, 26].
An HFSS, FEM-based, 3D full wave electromagnetic solver simulator by ANSYS as well as CST were used for the design of all designed antennas. Also, ADS was used to design the filters and filtennas in addition to the above electromagnetic simulators in this chapter.
This chapter describes one example for printed millimeter wave antenna implementations, illustrating specific and interesting particular solutions for their design and two shapes of single UWB antenna in radio frequency range. In addition, two examples of UWB filters and one example of UWB filtennas are introduced.
Classical antenna as reflector, lens, and horn type antennas have been used in millimeter-wave devices. But for low-cost, these antennas are commercial expensive devices, bulky, heavy and require complex feeding in an array system. In addition, they are very difficult to integrate with solid-state devices [6, 7, 8, 9]. However, the microstrip antennas (MPAs) are narrow bandwidth and are large size about half-wavelength structures.
Three different types of broad multi-band linearly and circularly polarized slot antennas (rectangular-, circular-, and triangular-shaped slots) for millimeter wave wireless communication applications [27] are shown in Figures 2 and 3, respectively. Proposed antenna consists of a slot radiator on the top metal layer and coupled to a rectangular dielectric resonator above the slot. The conventional microstrip-line-feed is used for different shapes of slot antennas. Final designed antennas were fabricated, and their characteristics were measured as reflection coefficient. The bandwidth of |S11| < −10 dB was extended from 19.5 up to 75 GHz. This band covers wireless MM-wave applications and wireless networks, and the WLAN, WPAN, and W-bands and most of 5th Generation mobile [28, 29, 30, 31, 32]. The average radiation efficiency and gain over the entire operating band are about 60% and 6 dBi, respectively [27]. Printed different shapes of slot antenna show small dimensions (Lg × Wg) cut at different shapes of slots on larger conductor and are centered above the microstrip-feed line. The microstrip feed line is composed of a straight section of length Lf. To improve the antenna reflection coefficient response, a square stub slot is added with the side length S = 0.6 mm for further improvement in the antenna impedance matching as shown in Figure 2. The width of the tuning line is equal to that of the 50 Ω microstrip line (Wf = 0.56 mm). The optimized dimensions of the proposed antennas are shown in Table 1.
Different shapes of slot antenna for LP (a) rectangular, (b) circular, (c) triangular, and (d) side view [27].
Shapes of slot antenna for circular polarization (a) rectangular, (b) circular, and (c) triangular [27].
Wg | Lg | W | Lr | Lftc | K | Lt | K2 | d1 | t |
7 | 10 | 4 | 2.9 | 11.6 | 0.25 | 4.5 | 0.6 | 0.7 | 4.5 |
Lfr | Lfc | Wf | S | Lt | C1 | C2 | K2 | d1 | t |
8.6 | 8.5 | 0.56 | 0.6 | 4 | 0.7 | 0.7 | 0.6 | 0.7 | 4.5 |
The optimized antennas (all dimensions in mm) [27].
The first shape of antenna design is either square with side length = 3.5 mm or rectangular slot with dimensions L = 2.9 mm and W = 4 mm as shown in Figure 2(a). The antenna reflection coefficient |S11| is shown in Figure 4(a). Figure 4(a) shows that two antenna designs were examined with the same start and end of the operating resonate frequency band from 19.5 to 75 GHz. The square slot shaped antenna has bandwidth discontinuities from 34 to 45 GHz. The second shape included in this study is the circular and elliptical shaped slot as shown in Figure 2(b). The ellipse slot major diameter is W = 4 mm, and different radius ratios were used including the design with a ratio equal to 1.35 as shown in Figure 4(b). The |S11| response shows that the optimal bandwidth at |S11| < −10 dB is obtained using a circular slot shape. The bandwidth extends from 21 to 80 GHz in this case, while for the case of the elliptical slot, the bandwidth starts from 28 GHz and has the same end at 80 GHz as shown in Figure 4(b). The triangular shape is the third slot shape used in this study, and it is compared to trapezoidal as an intermediate stage between a triangle and a rectangle slot as shown in Figure 2(c). Figure 4(c) shows that both of the stated shapes resonate at frequencies higher than the other previous shapes, as it starts from 29 GHz.
Antenna |S11| response with different slot shapes (a) rectangular, (b) circular, and (c) triangular [27].
Single-feed MPAs for CP are usually achieved by using traditional simple changes in the shape of the patch such as truncating corners, using nearly square or nearly circular patches, cutting a diagonal slot in the square or circular patches, protruding or inserting a pair of symmetric perturbation elements at the boundary of a circular patch [33, 34, 35]. However, this type produced narrow axial ratio (AR) bandwidth. The dual-fed and sequential array structure produced wider AR bandwidth, but this requires more complicated design and may occupy larger space. Slot antennas are expected to overcome the limited bandwidth problem as well as similar changes in the slot shape could be used to achieve CP.
Rectangular notches etched in the rectangular slot are used as a way to improve the 3 dB AR bandwidth as shown in Figure 5(a). The notch width C1 = C2 = 0.7 mm gives the best wideband AR. The comparison of AR between measured and simulated is also shown in Figure 5(a).
Axial ratio of the antennas with different slot shapes (a) rectangular, (b) circular, and (c) triangular [27].
In the proposed design, a square stub with dimension side K was added to the circular shaped slot to achieve the CP performance. The stub is added at a radial distance t from square stub. The locations of this stub set the AR bandwidth without degrading the antenna performance as shown in Figure 5(b). This figure shows that the stub with notch at a distance t = 3 mm gives the best performance AR bandwidth. However, AR bandwidth discontinuities appear from 40 to 55 GHz. The comparison results of AR values simulated and measured at t = 3 mm is shown in Figure 5(b). Finally, for the triangular slot shape, a truncated corner was used. To improve the AR bandwidth, an L-shaped strip was added at the other triangular corner with width 0.1 mm as shown in Figure 3. The bandwidth for simulated and measured AR values for the triangular CP antenna is shown in Figure 5(c). From previous shapes, it appears that rectangular shaped slot with notches gives wide axial ratio bandwidth without degrading the antenna bandwidth.
The proposed antenna designs were fabricated by using milling machine technology with 0.1 mm accuracy on Rogers 6035HTC substrate with a 0.25 dielectric thickness and 0.017-mm copper thickness. A 1.85-mm end launcher connector is used to measure the proposed antennas. The simulation reflection coefficient was verified by comparison with the experimental results of the antennas by using 37397C Anritsu vector network analyzer. Photos of the fabricated antennas are shown in Figures 6 and 7. The comparison between measured and simulated |S11| for linearly and circularly proposed antennas are shown in Figure 8(i) and (ii), respectively. The measurement and simulation result data are in a good agreement. Measured results ended at 65 GHz as it is the end-point of the network analyzer. The rectangular slot shape gives the best antenna performance for linearly and circularly polarized slots. These designs also have low profile, wide impedance bandwidth |S11| < −10 dB, and wide 3 dB axial ratio.
Photo of fabricated linear polarized slot antenna, (a) rectangular, (b) circular, and (c) triangular shaped [27].
Photo of fabricated circular polarized slot antenna, (a) rectangular, (b) circular, and (c) triangular shaped [27].
Measured and simulated |S11| of (i) linear polarized (a) rectangular, (b) circular, and (c) triangular slot shaped and (ii) circular polarized (a) rectangular, (b) circular, and (c) triangular slot shaped [27].
Planar microwave circuitry has generated attractive radiating structures with high gain, low weight, reliability, ease of manufacturing and integration such as the Vivaldi antennas [36, 37], and the tapered slot antenna [38] for UWB antennas. The printed planar log-periodic dipole (LPDA) is the most suitable solution microwave frequencies [39]. LPDAs have a lot of advantages, such as directive radiation pattern, linear polarization and low cross polarization ratio over a wide frequency range [5]. At the beginning, coaxial cable was used for feeding the printed LPDAs at the radio and the TV frequency bands; however, it was found that the performance became worse when frequency increases. Due to huge increase in data traffic, there is a requirement for wireless networks which support both data and voice transfer simultaneously for short-range wireless communication systems [1, 2].
This section presents a LPDA for UWB applications [39]. The antenna consists of cascading four U-shaped elements of different line lengths with balun circuit to improve the antenna impedance matching. The proposed antenna area dimensions are 50 × 50 mm2 built on FR4 substrate thickness 0.8 mm. Simulator HFSS is used for modeling the designed antenna. The pulse distortion is verified by measuring the proposed antenna performance with virtually steady group delay. The simulation and measurement results show that the designed antenna exhibits good impedance matching, stable radiation patterns throughout the whole operating frequency bands, acceptable gain and stable group delay over the entire operating band.
LPDA is UWB with the multiple resonance property; its bandwidth can be enhanced by increasing the number of the dipole elements [40, 41, 42]. Balanced structure, CPW fed antennas are very good candidates since the feed lines and the slots are on the same side of the substrate. There are many researches done to design LPDA but most of the published papers are not compact and their length is near from wavelength. The compact antenna dimensions, as shown in Figure 9, are 50 × 50 × 0.8 mm3. The proposed USLPDA antenna introduces USUWB with the multiple resonant property and compact size compared to earlier designs where UWB was realized using a rectangular slot [43], U-shaped dipole elements or stubs [38]. The bandwidth of this antenna at −10 dB reflection coefficient extends from 1.85 to 11 GHz which is wide enough to cover the FCC approved UWB in addition to wireless communications. The antenna exhibits good performance and can operate at wireless applications.
Layout of the proposed log periodic dipole antenna (semi-LPDA) [39].
The designed geometry antenna is shown in Figure 9; the antenna is composed of four different lengths with U-shaped stubs. The lengths and spacing of the elements of LPDA increase logarithmically from one end to the other. The design of the LPDA is used where a wide range of frequencies is needed while still having moderate gain and directionality. The simulator HFSS is used to validate and optimize by simulating the designed antenna. The designed antenna is built on a commercial FR4 substrate with dielectric constant εr = 4.6, and loss tangent tan δ = 0.02. The antenna is fed by a 50 Ω transmission line, which can be easily integrated with other microwave circuits printed on the same substrate. For designing procedure, the number of trial steps is needed, the scale-factor τ, spacing factor δ, and the number of the dipole elements N should be determined. Second, the length of the longest arm, which responses to the lowest resonance frequency f1, should be computed by following Eqs. (1)–(6) [39].
where λ1eff, Bo, N, and int i are the longest effective operating wavelength, the operating frequency, number of elements, and i is an integer that varies from 2 to 5, respectively. To improve the impedance, matching the balun circuit with suitable dimensions is used as shown in Figure 9. Table 2 introduces the dimensions of the proposed antenna [39].
Lsub | Lg | Wsub | Wg | W1 | W2 | W3 | S | g | W4 |
50 | 13.5 | 50 | 24 | 15.3 | 11.7 | 8.5 | 0.9 | 0.6 | 6 |
Lsep | L4 | Lfeed | L3 | L1 | L2 | Wf | K | P | d |
7.6 | 2.1 | 45 | 2.8 | 3.6 | 3 | 6 | 8.5 | 4.5 | 1000 |
Dimensions of the proposed antenna (dimensions in mm) [39].
The designed antenna is fabricated by using photolithographic technique, as shown in Figure 10, and parameter performance is measured. The simulated and measured input reflection coefficient of the antennas is in good agreement, as shown in Figure 10(b). The designed dipole impedance bandwidth at −10 dB of antenna extended from 1.85 to 11 GHz to cover most of the wireless applications and FCC UWB regulation. The antenna gain data are compared between measured and simulated results as shown in Figure 11. The designed LPDA antenna achieves simulated average gain 5.5 dBi, and the peak realized gain around 6.5 dBi at 2.7 GHz as shown in Figure 11(a). The measured results show very good agreement with simulated results and about ±3 dBi difference on average over the operating band. Wheeler cap method [44] can be used to calculate the antenna radiation efficiency. The average radiation efficiency is around 70% over the operating band as shown in Figure 11(b).
(a) GD simulated structures and (b) comparison between measured and simulated GD of LPDA [40].
(a) Fabricated LPDA antenna and (b) |S11|comparison between simulated and measured results [40].
Then the radiation efficiency measured result is done by using horn antenna to complete the designed antenna radiation efficiency measurement as shown in Figure 11(b). In the designed antenna, the radiator and the ground plane are contributing to radiation. For UWB applications, omnidirectional radiation pattern is an important requirement. At lower frequency band of operation, the pattern resembles a conventional dipole antenna, but at higher end of the UWB spectrum, some ripples are observed which are attributed to higher order modes. There are discrepancies observed at higher frequency band spectrum, which arises due to measurement setup.
For UWB applications, group delay is an important factor in communication systems, for example, medical applications systems, security systems, and satellite communication systems, which are used for transmitting. To avoid occurring of distortion, it is recommended that the spectrum is treated in the same manner, over the proposed bandwidth of frequencies. When GD ripples are large, they may cause unsatisfactory distortion in the signal of a transmitting radio system. So, in radio system designs, there are specifications for how much a GD may be accepted. In nonlinear systems, nonlinear distortion happens since the magnitude of frequency response is not constant, and the phase response is nonlinear. The phase distortion could be used to measure GD, the phase characteristics must have a linear slope so that the ratio is constant for all frequencies and this represents a constant GD [44]. To measure the GD between two antennas with spacing d = 1 m, the usual practice is to derive Q/ω for |S21| phase. However, it is desirable that the same antenna is used for transition and receiving. High GD variations occur due to the steep phase shift over frequency, which may cause unsatisfactory distortions in the signal. Figure 12 illustrates the simulated GD, and it can be noticed that the average group delay is about 1.5 × 10−9 second.
Comparison between simulated and measured results (a) gain and (b) radiation efficiency of LPDA [40].
This section presents designed steps to model an UWB monopole antenna. The designed antenna is composed of three different lengths of semi-circular shapes connected with circular disk and half circular modified ground plane. The designed antenna has an area equal 50 × 50 mm2 on a low cost FR4 substrate [45]. The antenna demonstrates impedance bandwidth of −10 dB extended from 1.5 to 11 GHz with discontinuous bandwidth at different interior operating bands. Two pairs of SRR as metamaterial structure cells are inserted closely located from feeding transmission line of the antenna to achieve good impedance matching over the entire band of operation and improve the antenna performance. The fundamental parameters of the antenna including reflection coefficient, gain, radiation pattern, and group delay are obtained, and they meet the acceptable UWB antenna standard. Simulator HFSS ver. 14 is used as full wave electromagnetic solver then the prototypes are fabricated and measured. Results show that the antenna is very suitable for the applications in UWB as well as wireless communication systems.
For use in UWB systems, printed monopole patch antenna (PMPA) is an extremely attractive candidate because of its wide impedance bandwidth, omnidirectional azimuthal radiation pattern, low profile, and ease of integration with active devices and fabrication [46]. The design equation for lower band edge frequency has been reported in the literature. Moreover, these antennas are analyzed by using the frequency domain characteristics like return loss, gain, radiation pattern, surface current distribution, and group delay. Different narrowband services like WLAN, WiMax, GSM, UMTS, Wi-Fi, WMTS, ISM, UNII, DECT, European Hiper LAN I, II, and UWB (3.1–10.6 GHz) applications [1] could be obtained by using single UWB antenna.
A SRR is one of the electrically smallest resonant elements. It has many applications ranging from compact filters to advanced metamaterials. SRR has also a significant importance in electrically small antennas [46]. Metamaterials are good candidate for enhancement of the performances of different antennas. There are varieties of SRR structures that have been reported in the literature like square, circular, triangular, omega, and labyrinth resonator [47].
The designed antenna structure is composed of three connected semicircular arc monopoles with circular patch fed by microstrip transmission line and modified half circular shaped ground plane as shown in Figure 13. The initial design is validated and optimized by simulating the proposed antenna. The proposed antenna is designed on FR4 substrate with height 1.6 mm, dielectric constant εr = 4.6, and loss tangent tan δ = 0.02. The antenna is fed by a 50-Ω transmission line (TL).
Evolution of the design steps of the proposed printed monopole. (a) First step, (b) second step, (c) third step, (d) fourth step, (e) fifth step, and (f) sixth step [45].
The main design parameter for UWB antenna is the lower frequency edge (fL) rather than the resonance frequency (fr) as in Eq. (7). The lower band edge frequency of the designed antenna is calculated approximately by equating their surface area with that of an equivalent cylindrical monopole antenna of the same height as given by [45]. If R1 is the height of the planar monopole antenna in cm, which is taken the same as that of an equivalent cylindrical monopole, and r in cm is the effective radius of the equivalent cylindrical monopole antenna, which is determined by equating area of the planar and cylindrical monopole antennas, then the lower band edge frequency is given as [45]:
where Lf is the length of the 50 Ω feed line in cm.
The design started with first semi arc 1800 with radius 25 mm and with 5 mm width modified ground plane Lg = 19 mm as shown in Figure 13(a), and related |S11| is shown in Figure 14. To improve the bandwidth, second semi sector with radius 17 mm and width 3.5 mm as shown in Figure 13(b) with the same previous dimensions is added to add second resonant as shown in Figure 14. Third sector with radius 7.5 mm and width 2.5 mm is added, and keeping previous dimensions the same as shown in Figure 13(c), a third resonance is achieved as shown in Figure 14. Another extension of the bandwidth is done by adding circular disk with radius 4 mm as shown in Figure 13(d), and related reflection coefficient is shown in Figure 14. Modified ground plane is used to improve the bandwidth with ellipse with major radius 25 mm and minor radius 15 mm, as shown in Figure 13(e), is suggested, and the related reflection is shown in Figure 14. The evolution of designing the proposed configuration is demonstrated in Figure 13(f), and their corresponding optimized dimensions are tabulated in Table 3. The antenna gain and radiation efficiency are also studied as shown in Figure 15. The proposed antenna with SRR achieves an average gain of 7.5 dBi, and the peak realized gain around 22.5 dBi at 7.5 GHz as shown in Figure 15(a). The designed antenna gain without SRR achieves an average gain around 5.5 dBi, while peak gain realized is 15 dBi at 8.5 and 10 GHz. The gain of the designed antenna is also measured, and there is good agreement between results especially at high frequency. The antenna radiation efficiency was simulated for both monopole antennas with and without SRR by using wheeler cap method [44]. The average radiation efficiency is around 70% over the operating bands for PMPA with SRR and around 65% without SRR as shown in Figure 15(b).
Lsub | Wsub | W1 | W2 | W3 | R1 | R2 |
50 | 50 | 5 | 3.5 | 2.5 | 25 | 17 |
R3 | Rs | S1 | S2 | Wf | Lf | Rg |
7.5 | 4 | 5 | 2.4 | 3 | 20 | 20 |
Dimensions of the proposed antenna (all dimensions in mm) [45].
Design procedures of the proposed antenna [45].
(a) Antenna gain versus frequency and (b) simulated radiation efficiency with and without SRR [45].
The prototype of the proposed antenna is shown in Figure 16. The performance parameters of the fabricated designed antennas are measured. The comparison results of simulated and measured input |S11| of the antennas are found to be in very good agreement, as shown in Figure 17. The −10 dB bandwidth of the designed monopole antenna with SRR extended from 1.5 to 11 GHz to cover most wireless application and FCC UWB regulation. It is fabricated by using photolithographic technique, and the measurements were carried out by using a Rohde & Schwarz ZVA67 vector network analyzer from 50 MHz to 67 GHz.
Fabricated antenna (a) upper layer without SRR (b) upper layer with SRR, and (c) bottom layer [45].
Simulated and measured results of (a) monopole without SRR and (b) monopole with SRR [45].
A filter is a two-port network that is used to control frequency response in a system. Filters can be classified into three main groups of active (requiring external power source), passive (no need for external power), and hybrid filters. Microwave systems are often involved with power conservation and noise control, and therefore, active filters are generally considered as last alternative. Passive filters are, however, further divided into lumped and distributed. The former consists of lumped components (including capacitors, inductors, resistors, and magnetic and electromechanical components), and the latter comprises a periodic conducting structure with various dielectric media. Inductors and capacitors form LC filters whereas resistors and capacitors form RC filters. Although resistors introduce loss to the circuit, they are generally used for broad banding (matching) purposes.
A compact UWB BPF with reconfigurable notch bands based on CRLH transmission line unit cell has been designed, simulated, and fabricated [48]. Two packages of software are used, namely, CST MWS and 3D EM commercial software HFSS version 13.0. The simulated and measured results are comparable. This filter has the advantage of very small size, and it also has four notched frequencies in its passband. The notched bands suppress the narrow-band services such as WLAN and WiMAX. One can control the center frequency of the notched band by varying the length L6 of the stub. The total area of the filter is 16.4 × 5 mm2. This small area makes it suitable for modern applications which need miniaturization.
The proposed filter is designed based on the filter described in Ref. [49] but with a new contribution which is notched controllable tunable four sharp rejection bands by adjusting the length of the coupling stub using diode switching matrix tools (instead of using PIN diodes).
Figure 18 shows the proposed microstrip UWB-BPF with tuned notched passband based on CRLH transmission-line unit cell. The optimized dimensions of the proposed filter are as shown in Table 4.
The proposed filter [48].
L1 | L2 | L3 | L4 | L5 | L7 | L8 | w1 | w2 |
9.3 | 3.4 | 4 | 5.2 | 7.2 | 5.3 | 3.3 | 0.3 | 0.2 |
g1 | g2 | g3 | g4 | g5 | g6 | g7 | g8 | g9 |
0.3 | 0.9 | 0.9 | 0.2 | 0.3 | 0.4 | 0.2 | 0.2 | 0.5 |
Optimized dimensions of the proposed filter (all dimensions are in mm) [48].
The dimension of the multi-mode section as shown in Figure 18 is 4.4 × 1.5 mm, the length of the shunted inductive line is 3.1 mm, and the overall dimension of the proposed filter is 16.4 × 5.0 mm. Based on the above description, the design procedure can be as follows:
The notched band depends on the coupling stub (L6) in the output section.
The notch frequency of the filter can be changed by adjusting the length of the coupling stub L6. As L6 increases, the center notch frequency decreases as shown in Table 5. The length L6 is controlled by using switching matrix equipment (mini circuit) where the character D refers to the diode.
Diode states | L6 (mm) | fnotch (GHz) |
---|---|---|
D1, D2, D3 (off) | 2 | 6.18 |
D1(on), D2, D3 (off) | 3.1 | 5.9 |
D1, D2(on), D3(off) | 4.2 | 5.7 |
D1, D2, D3 (on) | 5.3 | 5.5 |
(fnotch) against (L6) variation [48].
The filter was fabricated using a photolithographic technique on Rogers RT/Duroid 5880 with εr = 2.2, h = 0.787 mm, and tan δ = 0.0009. The photograph of the fabricated filter is shown in Figure 19. The measured and simulated S11 and S21 for different stub lengths are shown in Figure 20(a)–(d). The UWB bandwidth extends between 3.1 and 10.6 GHz. There are four notched frequencies for the different stub lengths (L6 = 2.2, 3.1, 4.2 and 5.3 mm). The overall dimension of the filter is 16.4 × 5 mm, which is considered very compact compared to other published designs with the same characteristics. Figure 20(a) shows the measured and simulated S11 and S21 with L6 = 2 mm and fnotch = 6.18 GHz. Figure 20(b) shows the measured and simulated S11 and S21 with L6 = 3.1 mm and fnotch = 5.9 GHz. Figure 20(c) shows the measured and simulated S11 and S21 with L6 = 4.2 mm and fnotch = 5.7 GHz. Figure 20(d) shows the measured and simulated S11 and S21 with L6 = 5.3 mm and fnotch = 5.5 GHz. Good agreement was found between the measured data and simulated results.
A photo for the fabricated filter [48].
The simulated and measured S11 and S21 for different L6 lengths, (a) L6 = 2.0 mm, (b) L6 = 3.1 mm, (c) L6 = 4.2 mm, and (d) L6 = 5.3 mm [48].
Ref. [50] proposed a suitable UWB to dual-band band-pass filter with defected ground structure DGS. This filter consists of four parts, namely, meandered inter digital coupled line sections, stepped impedance open stubs, coupled lines, and rectangular DGS. The filter is miniaturized and has a total area of 12.5 × 10 mm, Figure 21. This filter is fabricated on Duroid Teflon substrate with a dielectric constant of 2.2 and a dielectric height of 0.7874 mm. The UWB mode extends from 3.6 up to 10.6 GHz with attenuation greater than 20 dB up to 18 GHz (upper stopband). The dual passbands extended from 3.8 up to 18 GHz (upper stopband). The dual passbands extend from 3.8 up to 5 GHz and from 9.5 up to 10.8 GHz. The proposed filter suppresses WiMAX and X (military) band of satellite that extends from 7 up to 8 GHz. The filter is designed, fabricated, and measured. The mode of the filter is changed by using suitable matrix equipment [50].
The structure of the proposed filter [50].
DGS at input and output ports of the proposed filter produces two resonances at 7.5 and 9.6 GHz and improve the performance of proposed filter while an overall size reduction of 20% is obtained. The meander lines and stepped impedance open stub are also used to reduce the overall size. By adjusting the connection between the coupled lines in the center of the design, the center frequency and 3 dB frequency band can be easily adjusted. The proposed filter achieves UWB performance with good selectivity and low insertion loss in the passband from 3.6 to 10.5 GHz and good stopband from 10.6 to 18 GHz. Moreover, it achieves dual bands with good stopband from 5 to 9.5 GHz and from 10.8 to 18 GHz by using open circuit stub to suppress unwanted interference signals in the band of WLAN, WIMAX, and X (Military) band of satellite. All dimensions of the proposed filter are as follows: L1 = 3.75 mm, L2 = 1.95 mm, L3 = 1.8 mm, L4 = 7.5 mm, L5 = 2.1 mm, L6 = 1 mm, L7 = 5.65 mm, W1 = 0.2 mm, W2 = 0.5 mm, W3 = 0.15 mm, g1 = 0.2 mm, g2 = 0.2 mm, and g3 = 0.3 mm. The simulated S11 and S21 are shown in Figure 24.
Figure 22 shows the equivalent lumped circuit model of the proposed UWB BPF that is shown in Figure 21. The equivalent lumped circuit model results are obtained using circuit model tool of the Advanced Design System (ADS) 2017. The lumped element values are manually optimized by changing each element value, so that it can have good agreement with the simulated results obtained from the full wave simulator.
Equivalent lumped circuit model of the proposed UWB BPF shown in Figure 21 [50].
The whole equivalent circuit of the proposed filter can be divided into the following subsections: DGS part at input and output ports, interdigital coupled lines and stepped impedance open stub. As shown in the lumped element model (Figure 22), Rd1, Cd1, Ld1, Rd2, Cd2, and Ld2 represent the equivalent resistance, inductance, and capacitance of the defected ground structure (DGS) [51]. L5, C5, L6, and C6 represent the equivalent inductance and capacitance of the stepped impedance resonator (SIR). Interdigital coupled arm is represented by the series capacitance with parasitic inductance and resistance, and shunt capacitances [52] as shown in Figure 22. The S-parameters versus frequency response of EM simulation and circuit model are compared. There is a very good agreement between the simulated and equivalent lumped circuit model results.
Photolithographic technique was used to fabricate this filter on Teflon substrate (Duroid RT 5880) with physical properties of εr =2.2 and tan∂ = 0.0009, while the dielectric thickness is 0.7874 mm. Figure 23 shows a photograph for the fabricated filter for both sides (the front and back sides). The soldered wires shown in Figure 23 are used to connect the filter with diode switch matrix tool. The filters are measured using the vector network analyzer (N9928A FieldFox Handheld Microwave Vector Network Analyzer, 26.5 GHz) [50].
A photo for the fabricated filter [50].
Figure 24(a) shows the measured and simulated results of the proposed filter at ON state with frequency range from 1 to 20 GHz. It should be noted that the frequency range is extended up to 20 GHz in order to show that the out of band rejection is good, and the measured 3 dB passband of the proposed filter is between 3.6 and 10.6 GHz. Figure 24(b) shows the measured and simulated results of the proposed filter at OFF state, and the dual bands with 3 dB passbands extend from 3.8 to 5 GHz and from 9 to 10.8 GHz [50].
The simulated and measured S11 and S21 without O.C stub. (a) D1 and D2 ON state (with frequency range from 1 to 20 GHz) and (b) D1 and D2 OFF state [50].
Photos for the fabricated filter with open stub are shown in Figure 25. Figure 26(a) shows the measured and simulated results of the proposed filter with open stub at D1, D2 ON state, and D3 OFF with frequency range from 1 to 20 GHz. It should be noted that the out of band rejection is good, and the measured 3 dB passband of the proposed filter is between 3.6 and 10.6 GHz. Figure 24(b) shows the measured and simulated results of the proposed filter with open stub at D1, D2 OFF state and D3 ON, and the dual bands with 3 dB passbands extend from 3.8 to 5 GHz and from 9.5 to 10.8 GHz [50].
A photo for the fabricated filter of Figure 10.
The simulated and measured S11 and S21 with O.C stub. (a) D1, D2 ON, and D3 OFF, (b) D1, D2 OFF and D3 ON [50].
In general, the filtenna consists of a filter and antenna that are combined in one structure. The proposed filtenna operates at three bands of frequency (2.4, 5.5, and 28 GHz) to cover the 4G/5G communication system. It consists of three parts, namely, Franklin strip monopole antenna to cover 4G, WLAN, and WiMAX and a rectangular patch antenna to cover 5G band. The third part consists of a modified CMRC low-pass filter that exists between the two antenna parts to isolate the Franklin antenna from the rectangular patch antenna at 5G band. It also allows feeding the Franklin antenna at low frequency bands. The total size of the filtenna is 45 × 40 × 0.508 mm3 and fabricated on Teflon dielectric substrate (Roger 5880). The proposed filtenna has wide impedance bandwidth (15.8, 23.5, and 11.3%) and high gain (1.95, 3.76, 7.35 dBi) [53]. The proposed multiband filtenna is shown in Figure 27.
The proposed multiband filtenna (a) front view and (b) back view [53].
A modified compact microstrip resonance cell (CMRC) low-pass filter (LPF) using novel fractal patches was proposed in [54], see Figure 28. The fractal patches produce additional transmission zeros to the stop-band, while the open-ended stubs cause an extension in the stopband achieving a compact ultrawide and deep stopband filter with good selectivity and low insertion loss in the passband. The results show −10 dB bandwidth from 3.3 to 67 GHz with 181.5% relative stopband bandwidth. The 3-dB cutoff frequency is 2.85 GHz and less than 1.5 dB insertion loss in the passband and 0.55 GHz transaction from −3 to −20 dB and − 20 dB suppression from 3.5 to 67 GHz, so that the filter can be expected to suppress the unwanted harmonics and prevent inter-modulation with the new systems with high frequency operating bands. The filter has been designed on a Rogers 5880 substrate with a relative dielectric constant of 2.2, substrate thickness of 0.508 mm, and 0.0009 loss tangent. Figure 28 shows the proposed filter design, and it consists of two traditional triangle taper resonance cells in one side of the transverse connecting narrow width transmission line which has almost the same performance of the complete CMRC structure, while two different sizes circular fractal patches are present on the other half. Each fractal consists of main circular patch and additional small circular patches at edges. The two fractals act as a dual behavior resonator to have additional transmission zeros in the stopband. Each fractal is resonating at certain frequency in addition with enhancing the low suppression bands of the entire stop-band. Also, four open ended stubs are used to extend the stopband by adding new transmission zeros without increasing the circuit size. The main dimensions are given in Table 6, all dimensions in millimeter.
The design of proposed low-pass filter [53].
Parameter | Cl | L | W | Cd1 | Cd2 | a11 | a12 | a13 | a21 |
---|---|---|---|---|---|---|---|---|---|
Value (mm) | 17.6 | 2.4 | 0.15 | 7.6 | 2.1 | 1.35 | 0.2 | 0.05 | 2.9 |
Parameter | a22 | a23 | X1 | X2 | Y1 | Y2 | T1 | T2 | W50 |
Value (mm) | 0.15 | 0.05 | 2.4 | 1.8 | 0.8 | 0.8 | 3.2 | 3.8 | 1.6 |
Circuit dimensional parameters [53].
Part C of the multiband rectenna system is a Franklin strip monopole dual-band antenna to be used to cover Bluetooth at 2.4 GHz, 4G, LTE bands at 2.3, 2.5 and 2.6 GHz, WiMAX at 2.5 and 5.5GHz, WLAN at 2.4, 5.2 GHz [53]. The geometry of the antenna is shown in Figure 29. The antenna has a rectangular stub on a curved partial ground. The length of the bending strip is about one-half of the guide wavelength at its first resonance frequency. The meander radius and the length of the Franklin strip are mainly determining the two resonance frequencies of the antenna, while the rectangular stub with a length of a quarter wavelength and curved ground have been used to increase the bandwidth of the upper band (5.5 GHz). The L-C equivalent circuit of the Franklin monopole antenna is shown in Figure 30. The dimensions of the antenna are shown in Table 7, while the equivalent circuit parameters are shown in Table 8.
The design of the proposed Franklin strip monopole antenna (a) front and (b) back [53].
The equivalent LC circuit of the proposed 4G Franklin monopole antenna,
Lsub | Wsub | Lf | Lg | L | d |
35 | 45 | 18.5 | 14.9 | 5.1 | 9.9 |
R | a | Wf | Ws | Tf | Ts |
1 | 1.1 | 1.6 | 1.5 | 13.4 | 14.9 |
Dimensions of the proposed Franklin antenna (all dimensions in mm) [53].
CTL | LTL | Cp | Lp | Rp | CS | LS | Ck | Ccs |
---|---|---|---|---|---|---|---|---|
2 | 1.7 | 2606 | 0.0007 | 0.05 | 0.1 | 0.05 | 18.4 | 42.4 |
Franklin monopole antenna equivalent circuit parameters [53].
C in Pico Farad, L in Nano Henry and R in ohm.
The first part of the rectenna (part A) consists of a rectangular patch antenna with inset feed for matching and four CSRRs (complementary split ring resonator) in the other side (ground plane). This antenna covers the 5G range of frequency (28 GHz). This shape is chosen due to its simplicity and can be placed in the Franklin feeding line. Figure 31 shows the geometry of the rectangular patch with inset feeding. The final dimensions of the antenna after using optimization techniques of the CST simulator are introduced in Table 9. The L-C equivalent circuit of this antenna with CSRRs is shown in Figure 32. The rectangular patch is represented as two radiating slots each one represented by a parallel R L C circuit, while each one of the four CSRRs represented by parallel LC and the electric coupling between the CSRRs on the ground and radiating patch on the top side is represented by capacitor Ck and the magnetic coupling represented by the mutual inductance offered by the ADS software. All lumped element values are listed in Table 10 [53].
The design of 5G rectangular patch antenna. (a) Front view, (b) back view, and (c) the dimensions of the CSRR [53].
(a) The equivalent LC circuit of the proposed 5G rectangular patch antenna [53].
Lsub | Wsub | Lf | Wf | yo | Lp | Wp |
11 | 7.5 | 6 | 1.6 | 1 | 2.95 | 3.45 |
Sx | Sy | a | b | c | e | f |
1.15 | 1.55 | 1.85 | 1.55 | 1.25 | 0.8 | 0.3 |
5G antenna dimensional parameters (all dimensions in mm).
CTL | LTL | Cp1 | Lp1 | Rp1 | Cp2 | Lp2 | Rp2 | Cc | Lc |
---|---|---|---|---|---|---|---|---|---|
17.5 | 0.16 | 1.06 | 0.26 | 93.6 | 10−12 | 0.02 | 54.9 | 0.79 | 1.2 |
5G antenna equivalent circuit parameters.
C in Pico farad, L in Nano Henry and R in ohms.
The equivalent circuit model simulation results of the filtenna system shown in Figure 27 is determined by merging the three parts component’s equivalent circuit, namely, 5G rectangular patch antenna, modified CMRC low-pass filter, and 4G Franklin monopole antenna extract using ADS software [53]. The filtenna was fabricated using the photolithographic technique.
Figure 32 shows a photo of the fabricated rectenna. The simulated and measured reflection coefficient is shown in Figure 33. The measured results show that the filtenna has −10 dB impedance of the first band from 2.16 to 2.53GHz, the second band is from 4.58 to 5.8GHz, and the third band is from 26.8 GHz to 30 GHz. The simulated and measured gains are shown in Figure 34. The first and second bands have peak measured gain level of 1.95 and 3.76 dBi, respectively. The third band achieves 7.35 dBi peak simulated gain level [53].
Photograph of fabricated antenna. (a) Front view, and (b) back view and (c) the measured and simulated reflection coefficient [53].
Simulated and measured gain at: (a) first band, (b) second band, and (c) simulated gain using CST and HFSS simulators for the third band [53].
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