Optoelectronic Key Elements for Polymeric Fiber Transmission Systems

2.1 Transmission properties of 1 mm PMMA SI-POF

The 1 mm PMMA SI-POF is the best known and by far the most widely employed type of POF. It is made of 980 μm diameter PMMA core surrounded by a thin cladding (10 μm) made of fluorinated polymer. The typical spectral attenuation of SI-POF is shown in Figure 3. The fiber supports operation in the visible spectrum from 400 to 700 nm. The lower wavelength bound is determined by the degradation of the PMMA compound with prolonged exposure to the ultraviolet (UV) wavelengths shown in [11] and in [12]. The attenuation value of around 400–450 dB/km, which still allows operation over shorter link lengths (<20 m), sets the upper wavelength bound.

Two intrinsic loss mechanisms contribute to the raise of attenuation at shorter and particularly UV wavelengths. The electronic transitions due to the absorption of light in the polymer compound cause absorption peaks in the UV region. However, their absorption tails extend through the visible spectrum affecting the POF attenuation [13]. The dependence of the attenuation coefficient of electronic transitions αe[dB/km] on the wavelength for PMMA is given by [14]:

The second loss mechanism is the Rayleigh scattering. It is caused by the structural irregularities in the polymer compound that are much smaller than the wavelength of light (order of one tenth of wavelength or less). The effect of scattering becomes more pronounced as the wavelength decreases since the scattering attenuation coefficient αs[dB/km] is inversely proportional to the fourth power of the wavelength shown by Kaino [15]:

In the infrared region, the attenuation significantly increases due to the intrinsic absorption losses caused by vibrations of the molecular C-H bonds (total of eight per MMA monomer). The higher overtones of the C-H bond vibrations also extend in the visible spectrum. The seventh overtone at 549 nm and particularly the sixth and the fifth overtone at 627 and 736 nm, respectively, cause pronounced absorption peaks and wide absorption bands, predominantly determining the level of attenuation in the red spectral range shown by Emslie [16] and by Groh [17].

The contributions of the intrinsic loss mechanisms to the overall attenuation of SI-POF are also shown in Figure 3. The wavelength regions where the fiber exhibits low attenuation are called attenuation windows. The SI-POF has four attenuation windows. Those are blue, green, yellow, and red windows, with the absolute attenuation minimum of approx. 62 dB/km at around 568 nm (yellow window). The parameters of the attenuation windows are listed in Table 2.

The mean refractive index of SI-POF core material in the visible spectrum is ncore=1.492, whereas the refractive index of cladding is nclad=1.412. Due to the big difference in refractive indices of core and cladding, the numerical aperture (NA)

has the value of 0.482 (usually rounded to 0.5). The corresponding maximum acceptance angle of the fiber is 30°. The large core radius acore= 490 μm combined with the high NA results in the normalized frequency V

of 2698 at 550 nm, which is far above the limit V = 2.405 below which a fiber is in single-mode operation. The number of modes Nmodpropagating through SI-POF can be approximated as

corresponding to 3.64 million modes at 550 nm. Due to the significant path difference between lower- and higher-order modes, propagating respectively at smaller and larger angles relative to the optical axis, the strong intermodal dispersion is inherent to SI-POF. In the time domain, it is manifested as pulse broadening, thus introducing the inter-symbol interference (ISI). In the frequency domain, the intermodal dispersion results in a low-pass frequency response, constraining the bandwidth-length product of SI-POF to around 50 MHz × 100 m shown by Ziemann et al. [18].

3.1 Data transmission techniques for increasing the channel capacity

Two data transmission techniques were used to overcome the bandwidth limitation of a POF WDM channel, which is primarily caused by the intermodal dispersion of SI-POF. Those were:

1. Non-return-to-zero (NRZ) modulation in combination with electronic dispersion compensation, in particular feed-forward equalization (FFE)

2. Discrete multitone (DMT) modulation

In the single-channel POF systems with intensity modulation and direct detection (IM/DD), those are well-known techniques for increasing the channel capacity. The next two subsections briefly introduce the two techniques: non-return-to-zero modulation and feed-forward equalization.

According to the Nyquist theorem for two-level signaling, the maximum bit rate (in bits per second) for a noiseless channel of the bandwidth Bequals 2B. Since the real systems encounter noise, this theoretical limit cannot be achieved. Therefore, for 100 m SI-POF link, the data rates less than 100 Mb/s are possible. If the data rate further increases, the eye diagram of the received signal becomes partially or completely closed due to the combination of the ISI and additive noise (introduced, e.g., by receiver’s transimpedance amplifier).

The equalization techniques are used to open the eye diagram at the receiver for clock and data recovery (CDR). In its principle, the equalizer compensates for ISI, which is deterministic (unlike the random noise) and determined by the low-pass frequency response of a POF channel. In the work a simple linear FFE equalization technique was employed to correct the distorted signal waveforms at the receiver.

An FFE equalizer is realized as a discrete-time finite impulse response filter with adjustable coefficients. The output of the equalizer is obtained as the weighted sum of the delayed samples of the input signal as

where Vetis the equalized voltage sample at the time t, ckis the equalizer coefficient (weighting factor), TDis the tap delay, and Vrt−kTDis the ktimes delayed uncorrected received voltage sample. A block diagram of the FFE equalizer is shown in Figure 5.

Through its coefficients the equalizer may synthesize a transfer function corresponding to the inverted channel frequency response, thus eliminating the ISI. In a noisy POF channel, and due to inverting the channel frequency response, this would lead to great noise amplification at higher frequencies where the channel frequency response is small in magnitude. Typically, to minimize the probability of the decision error, the weighting factors are calculated to minimize the noise power at the cost of a certain amount of residual ISI after equalization. The descriptions of different algorithms for optimizing the equalizer coefficients can be found in Loquai et al. [23].

The BER performance of the system was estimated based on the Q-factor of the equalized eye diagram (Figure 6)

where V1and V0are the mean values and σ1and σ0are the standard deviations of the equalized signal voltages associated with logic 1 and 0 levels, respectively, as shown in Figure 6. The corresponding BER was obtained as

where erfcx=2/π∫x∞exp−τ2dτis the complementary error function.

3.2 DMT: discrete multitone modulation

To provide high spectral efficiency of the signals transmitted within POF WDM channels, a DMT modulation technique was used. The DMT is a multicarrier modulation format and represents a baseband version of a better-known orthogonal frequency division multiplexing (OFDM). Unlike the OFDM, which is used in wireless communication systems such as wireless local area networks (WLAN), the DMT is widely employed as enabling technology for digital subscriber lines (DSL), e.g., asymmetric DSL (ADSL) and very high DSL (VDSL). The DMT-based transmission was also shown to be very beneficial for SI-POF communication by Joncic et al. [24], Diaz et al. [25], and Vinogradov et al. [10].

The DMT technique slices the frequency-selective channel into a large number of subchannels that can be considered to have a flat frequency response. Each subchannel is then used for transmission of a passband signal with quadrature amplitude modulation (QAM). The simultaneous transmission of the low-speed parallel streams reduces the influence of the ISI. Another important property of DMT is that it adapts the signal parameters (QAM size and power in each subchannel) to the characteristic of the communication channel.

The principle of the DMT transmission over an optical IM/DD channel is shown in Figure 7. A high-speed serial data stream is first divided into Nparallel lower-speed streams, where Ncorresponds to the number of subcarriers. Every Mbits in each stream are grouped together and mapped into a complex value corresponding to a point of the QAM constellation with 2Msize. The modulation of the complex values onto Ndifferent subcarrier frequencies is based on the inverse fast Fourier transform (IFFT). To obtain real-valued time domain signal samples at the output of the IFFT block, a 2N-point IFFE has to be carried out. Thereby, the second half of 2NIFFT inputs must be complex conjugate of the first half and symmetric around the middle of the input vector, a property referred to as a Hermitian symmetry. The real-valued samples of the DMT time signal are then parallel-to-serial converted, and a cyclic prefix (CP) is added as a guard interval at the beginning of each DMT frame to resist to inter-frame interference occurring in a band-limited channel. After digital-to-analog conversion, a bipolar continuous-time DMT frame is produced. It is then DC-biased to modulate the intensity of an electrical-to-optical converter.

At the receiver, the DMT waveform is direct-detected, analog-to-digital and serial-to-parallel converted, and demodulated using 2N-point fast Fourier transform (FFT). The first Ncomplex outputs of the FFT block are QAM-decoded and parallel-to-serial converted to result in an output serial data stream.

4.1 Operating principle and measurement results for a four-channel demultiplexer

By extending the channel count to four, it was possible to investigate two different demultiplexer configurations. Those were:

• Serial configuration

• Two-stage configuration

The principle of operation of a four-channel demultiplexer with serial configuration is shown in Figure 8. In this configuration the dichroic mirrors were cascaded such that each mirror (except the last one) demultiplexed a single-wavelength channel while passing all other wavelengths.

For practical realization 425, 505, and 567 nm cutoff long-pass dichroic mirrors were cascaded so that the interference filters centered at 405, 450, 525, and 650 nm could be implemented in the output ports 1–4, respectively. The corresponding transfer function and the basic parameters of the demultiplexer are shown in Figure 9 and Table 3, respectively. The factors contributing to high IL in the output ports 3 and 4 are discussed in the next subchapter.

The principle of operation of a four-channel demultiplexer with two-stage configuration is shown in Figure 10. The first stage of the demultiplexer, represented by a dichroic mirror that follows directly after the collimating lens, splits the incident spectrum into two spectral bands. The separation of the individual wavelength channels was then performed within the second stage of the demultiplexer. The corresponding laboratory setup is shown in Figure 11.

The practical realization was carried out with 505 nm cutoff long-pass dichroic mirror in the first stage of the demultiplexer. It reflected the lower spectral band so that 425 nm cutoff dichroic mirror was used in the second stage to demultiplex the signals for the output ports 1 and 2 in which 405 and 450 nm filters were employed, respectively. The upper spectral band transmitted by 505 nm mirror was demultiplexed in the second stage by 567 nm dichroic mirror. The filters centered at 525 and 650 nm were used in the output ports 3 and 4, respectively.

The corresponding transfer function is shown in Figure 12. The basic parameters of the demultiplexer are given in Table 4. The measurement results for the four-channel demultiplexer with two-stage configuration were presented at the International Conference on Plastic Optical Fibers (ICPOF) 2013 [26]. To comply with all other measurements shown in this chapter, which were performed 2 years thereafter, the demultiplexer setup was assembled and characterized again. While preserving the same principal behavior of the spectral response, the minimum IL in the output ports 1 to 4 was 6.15, 5.44, 4.21, and 3.85 dB, respectively. Those were by 0.49, 0.89, 0.74, and 0.66 dB higher values than those reported by Appelt et al. [26].

6.1 Experimental setup

The data transmission setup is shown in Figure 13. It comprised an Agilent N4903A bit error rate tester (BERT), four butt-coupled edge-emitting laser diodes, multiplexing POF coupler, 10 m SI-POF link, interference-based POF demultiplexer, optical receivers, and Agilent 86100B sampling oscilloscope.

To provide precise temperature control, prevent possible damage from overheating, and extend the lifetime, the laser diodes were mounted in Thorlabs TCLDM temperature-controlled laser diode mounts. The temperature of an integrated temperature control (TEC) element of the mount was adjusted to +15°C. Only for OSRAM samples the cooling at +10°C was used to provide better stability of the optical output power. Both the temperature of the TEC element and the bias current were controlled over a Thorlabs ITC8022 module. Four of those modules were installed in a Thorlabs PRO8000 modular chassis for the simultaneous control of four operating diodes (see Figure 15). To maximize the coupling efficiency from the laser diode into the fiber, a butt-coupling unit based on an xy-translator (0.25 mm pitch of the adjustment screws) was utilized. The translator was mounted on the TCLDM9 mount over a 4-rod construction system, as shown in Figure 15. To combine the optical signals with different wavelengths onto the SI-POF link, a Comcore 4 × 1 fused POF coupler was used (see Table 5).

6.2 Measurement results and discussion

6.2.1 Gb/s transmission over 10 m SI-POF

The laser diodes providing the signals at four different wavelengths operated at 405 (DL-5146-101S), 450, 515, and 660 nm. Each diode was inserted into a laser socket of the TCLDM mount, which was also equipped with an internal 500 MHz bandwidth bias tee and had separate inputs for the bias and modulating current. The respective data rates achieved in the individual WDM channels were 0.5, 0.5, 0.7, and 0.8 Gb/s. The corresponding eye diagrams are represented in Figure 14. The effect of pulse shaping due to the low-pass characteristic of the fiber can clearly be recognized, e.g., in 405 nm channel.

The irregular signal trajectories, e.g., in the eye diagrams of 515 and 660 nm channels, indicate the presence of nonlinearities in the electrical domain (presumably introduced by the bias tee circuit). No BER measurement data were saved. However, due to the eye diagrams still opened wide enough, it can be reasonably assumed that the corresponding BERs were below the FEC threshold of 10–3, allowing for the error correction.

High optical isolation of the demultiplexer provided very low optical crosstalk between the WDM channels. The crosstalk of ≤35 dB, coming from 450 nm channel, was detected in 515 nm channel. In all other channels, the crosstalk lower than −45 dB was detected with a Melles Griot 13 PDH 005 integrating sphere. Considering the amplitude levels of the recorded eye diagrams and low interchannel crosstalk, no reduction in the SNR of the received signals could be assumed if the laser diodes were modulated simultaneously. Therefore, it can be stated that an aggregate bit rate of 2.5 Gb/s could be transmitted over 10 m SI-POF with four simultaneously active channels and no interchannel errors. The corresponding information rate after deduction of 7% FEC overhead would be 2.33 Gb/s.

6.3 5 Gb/s transmission over 10 m SI-POF

In the next experiment, 405 (DL-5146-101S), 450, 515, and 639 nm laser diodes were used as WDM optical sources. To directly modulate the diodes with a higher modulation bandwidth, each diode was soldered to a 50 ohm SMA formable coax cable (Figure 14), which was connected to the output of an external bias tee with 6 GHz bandwidth and 0.1 MHz low cutoff frequency.

Due to the low impedance of the laser diodes (typically 2–5 ohms), a severe impedance mismatch was present. However, by using higher power of the modulating signals, the mismatch could be compensated. For mounting the diodes into the mounts, copper retainer rings for both TO-56 and TO-38 packages were fabricated (Figure 15). For better thermal conductivity between the TEC element of the mount, the retainer ring, and the laser diode housing, a heat-conductive paste was applied on the contact surfaces.

The bit rates in the individual channels were 1.25 Gb/s (405 nm channel), 1.05 Gb/s (450 nm channel), 1.25 Gb/s (515 nm channel), and 1.45 Gb/s (639 nm channel). The transmission parameters for the individual channels are listed in Table 6. In contrast to the previous measurement, no signal nonlinearities were present, and the maximum achievable data rates were limited by the ISI.

Taking into account the amplitude levels of the recorded eye diagrams and low interchannel crosstalk, like in the previous experiment, no power penalty due to crosstalk could be assumed if the laser diodes were modulated simultaneously. Therefore, it can be stated that 5 Gb/s transmission could be realized over 10 m SI-POF link at the BER <10^–4 with four simultaneously active channels and no interchannel errors. Using a standard Reed-Solomon (255,247) FEC with 3.2% redundancy, 4.84 Gb/s transmission could be achieved at the BER<10^–9 [28].

6.4 WDM transmission employing NRZ modulation and FFE equalization

This subchapter shows the improved experimental setup and gives the measurement results of the simultaneous four-channel NRZ transmission over 50 m SI-POF. To mitigate the effects of ISI, the FFE equalization was implemented at the receiver side. The experimental setup is shown in Figure 16. It comprised a four-channel Agilent M8190A arbitrary waveform generator (AWG), four butt-coupled edge-emitting laser diodes, four-legged multiplexing POF bundle, SI-POF link of two different lengths, interference-based POF demultiplexer, Graviton SPD-2 receiver, and four-channel Agilent DSA91604A real-time oscilloscope with built-in software for digital signal processing. A photo of the general setup for investigating four-channel high-speed POF WDM transmission is shown in Figure 17.

To multiplex the signals from four laser diodes onto the SI-POF link, a four-legged POF bundle was used. A multiplexing interface is formed by positioning the fiber bundle against 1 mm SI-POF.

For realization of the bundle, an Asahi KASEI DB-400 PMMA SI-POF with 400 μm cladding diameter and NA = 0.5 was used. Four 60-cm-long fibers were terminated at the input side with 400 μm FC connectors. The opposite ends of the fibers were joined together and glued inside 970 μm FC connector to form the fiber bundle (Figure 18). As illustrated in Figure 18, an FC connector-mating sleeve was used to bring together and align the bundle and the input of the SI-POF link, thereby forming the multiplexing interface. An index-matching gel was applied between the connectors to reduce the losses.

The described multiplexing solution was first shown in [29]. Shortly before, the patent application for an optical POF multiplexer based on a multi-legged POF bundle, which referred to arbitrary channel counts and fiber diameters, was submitted to the German Patent and Trade Mark Office (DPMA) under number DE 102013 020236.1. A similar approach was later adopted in [30, 31] to realize the low loss seven-legged and three-legged multiplexers, respectively.

The AWG simultaneously generated four independent NRZ data streams (Figures 16 and 19) based on 27-1 PRBS with the maximum sampling rate. The received electrical signals were acquired by the real-time oscilloscope with 8-bit vertical resolution and oversampling. The digital receiver equalization was carried out in the offline mode. For that purpose the oscilloscope’s built-in Serial Data Equalization software was used [32].

To prevent the equalizer from amplifying the noise components at higher frequencies where the energy content of useful signal was low, the bandwidth of the oscilloscope was set to the value in GHz corresponding to one half of the data rate in Gb/s. A phase-locked loop was used to extract the clock from the equalized data.

The eye diagram of an equalized waveform was displayed on the oscilloscope’s screen for further analysis. The oscilloscope’s built-in software EZJIT Complete was used to estimate the corresponding Q-factor [33]. Thereby, only a small time window (2% of the unit interval) in the middle of the equalized eye diagram was taken into consideration. The BER value was then calculated using Eq. (5). Table 7 shows the optical power losses of the four used WDM channels.

6.5 7.8 Gb/s transmission over 50 m SI-POF

The WDM channels based on 405 nm (DL-7146-101S), 450, 515, and 639 nm laser diodes were employed in this transmission experiment. To maximize the modulating signal amplitude and thereby improve the SNR of the received signal, the optical output power of each laser diode was adjusted to its maximal possible value. For 405, 450, and 515 nm devices, the operating point was set to comply with the upper limit of the receiver’s dynamic range. The 639 nm device was driven with the maximum recommended forward current. The respective used bias currents were 70, 40, 61, and 43 mA. The optical powers coupled into the SI-POF link and the received optical powers measured after the demultiplexer are given in Table 7. All diodes were driven in their linear lasing region. The 1 Vpp output amplitude of the AWG was sufficient to modulate 450, 515, and 639 nm laser diodes with the modulation index of approx. 0.9. The signal in 405 nm channel was additionally amplified to achieve the same modulation index. For the amplification, a MERA-556+ wideband amplifier (20.5 dB gain at 0.1 GHz) was used in combination with 10 dB attenuator to avoid amplifier nonlinearities (18 dBm output power at 1 dB compression point at 0.1 GHz).

The maximum transmission rates achieved in the individual channels were 1.7 Gb/s (405 nm channel), 1.9 Gb/s (450 nm channel), 2.2 Gb/s (515 nm channel), and 2 Gb/s (639 nm channel). Thereby, six FFE taps with the tap delay equal to one half of the corresponding bit period were used in each of the channels. The transmission parameters for the individual channels are listed in Table 8. The resulting eye diagrams are represented in Figure 19. A total of 7.8 Gb/s were transmitted over 50 m SI-POF at the BER < 10^–5. After deduction of 3.2% redundant bits required for Reed-Solomon (255,247) FEC, a net bit rate of 7.56 Gb/s was obtained. Compared to the record capacity of a single-channel system over the same fiber length used by Vinogradov et al. [10], an improvement of 1.67 Gb/s was achieved.