Numerical Simulations of Radiation and Scattering Characteristics of Dipole and LOOP Antennas

© 2012 Yurtsev and Ptashinsky, licensee InTech. This is an open access chapter distributed under the terms of the Creative Commons Attribution License (http://creativecommons.org/licenses/by/3.0), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited. Numerical Simulations of Radiation and Scattering Characteristics of Dipole and LOOP Antennas


Introduction
Dipole and loop antennas are used in communication lines, in radar and navigation systems, TV and other fields of radio-engineering. Structures of antennas depend on their bandwidth. In VHF, UHF, L and S ranges, these antennas are made, as a rule, of conductors with a radius o A   (λ -wavelength in the air). Further, we will study wire dipole and loop antennas.
Dipole and monopole antennas [Balanis, 1997;Markov, 1960], a Yagi-Uda antenna [Volakis, 2007;Aisenberg, Jampolsky and Terjoshin, 1977), bicone and discone antennas [Drabkin and Zuzenko, 1961], loop antennas consisting of one or two loops located in one plane [Rothammels, 1995] are well known. The length of a loop perimeter is L   . This ensures tuning of the antenna in resonance (a reactive part of input resistance is 0) r X  . To reduce sizes of dipole antennas, conductor with surface reactive resistance is used. Such a conductor can be implemented, for example, in the form of a cylindrical helix with a radius A   [Yurtsev, Runov and Kazarin, 1974]. To tune antennas in resonance, reactive resistances are used. Linear arrays of dipole antennas with series excitation are described briefly [Rothammels, 1995].
In communication nodes and antenna arrays, dipole and loop antennas function in the conditions of strong coupling. Therefore, one of the problems of numerical simulation is the research into the influence of interaction on the electrical characteristics of antennas.
Scattering characteristics of antennas are a strong factor that facilitates radar reconnaissance. These characteristics of loop and dipole antennas have hardly been studied in the literature.
In the literature, there is no enough information or there is no information at all about a number of questions related to research into and the versions of construction of dipole and loop antenna. The following is related to these problems: -optimization of dipole and loop antennas of complex shape according to various criteria (input resistance, directional pattern, bandness); Yagi-Uda multielement antennas, discone and bicone antennas, loop antennas with a quasi-isotropic directional pattern in the E-plane, a combination of loop and dipole antennas, all these antennas are attributed to complex antennas; -arrays of dipole and loop antennas with series excitation; -interaction of dipole and loop antennas in the near-field region; -scattering characteristics of dipole and loop antennas.
The above specified questions are the contents of the present chapter.

Method of analysis and mathematical model of wire antenna 2.1. Integral and matrix equations to determine current in arbitrary thin conductor
Wire antennas are simulated numerically by the method of integral equations. In the present chapter, we have used Pockington's integral equation [Mittra, 1973]. The integral equation is solved by Galerkin method (method of the moments) [Harrington, 1968;Mittra, 1973;Fletcher, 1984]. To integrate the integral equation to a system of linear algebraic equations, impulse functions have been used further as basis and weight functions. In this case, the basic part of each matrix coefficient of this system is defined by an exact analytical expression. That ensures the enhancement of accuracy of solution of the integral equation and reduces this problem time.
According to [Mittra, 1973] The right side depends on distribution of the extraneous field that excites the conductor: is the wave number of free space,  is the wavelength of the field that excites the conductor; os E  is the vector of the extraneous electric field that excites the conductor.
Green's function is defined by the expression: The remaining quantities included in (1) where M is the number of segments, into which the whole conductor of length L is divided; m is the number of a segment, in which point Q (source point) is located ( fig. 1 The right side of the system of equations (5) according to (3) turns out to be equal to: The weight function is distinct from zero and is equal to unity only within a segment with number n. Therefore, it follows from (15) that: where n E  is the component of an extraneous field, tangent to a conductor in a segment with number n; n U , n  are the amplitude and the phase of an exciting voltage in a segment with number n.
The further transformation into (16) is different for radiation and scattering problems.

Radiation problem
When solving a radiation problem, the extraneous field is considered to be distinct from zero only in the space of excitation and constant as for its amplitude and phase within the space. In this case, it follows from (16) that: where U is the excitation voltage; 0 n U  only in those segments, which are within the space of excitation. It means that, in the system of equations (5), the right side is given in accordance with (17) in the equations, the numbers of which correspond to the numbers of segments within the space of excitation.
As a result of numerical solution of the system of linear algebraic equations (5), the current in each segment m I is found. The amplitude and the initial phase of the current within each segment is considered to be constant. According to the current distribution in a conductor and the given voltage U, the input resistance inp Numerical Simulation -From Theory to Industry 162 inp inp U, I are the input voltage and the input current. The input voltage is the given voltage U in the space of excitation. The input current is the current at the ends of the segment (segments) that fills (fill) the space of excitation in a conductor model. The situation is explicated by fig. 2 that shows a part of a conductor of an antenna and some segments.
According to the found current distribution in the conductor, the geometry of the conductor, it is easy to determine numerically the conductor field in space in any zone, and further, the basic characteristics and parametres of a wire antenna: its directional pattern, phase diagram, polarization pattern, directivity factor. These questions compose the contents of an exterior problem of the theory, it will be studied in more detail below.

Scattering problem
The mathematical model of a wire antenna in the scattering mode includes the above relationships that determine current in scattering conductors, and the expression that defines the right side of the matrix equation (3). We will study (3) for the case of excitation of a conductor by a plane wave falling from the given direction. The problem is explicated by fig. 3.

U Сonductor
Model of conductor

Space of excitation
If the antenna is irradiated by a plane electromagnetic wave, the amplitude of the electric field is the same at all the points of the antenna and equal to io E . In this case, in the expression (20) fig. 3, angle i  is calculated from the Х axis to the Y axis).
The phase () p Ul in a segment with number m depends on the segment position in space, and is determined by the expression: ( cos cos sin sin cos ) where , , mm m XYZare the coordinates of the centre of the segment with number m.

Radiation (scattered) field of thin conductor
The field at an arbitrary point of space P is determined according to the found current distribution () q Il in a conductor. The positon of the point Р is set by spherical coordinates R, fig. 4. To calculate the electric field at the point Р, the vector-potential method has been used [Stratton, 1941;Volakis, 2007]: where pq R is the distance from point of observation Р to point Q on the antenna (see fig. 1). considered to be constant within each segment with number m . Therefore, the integral (25) is transformed into the sum of fields of M segments: ; m R is the distance from a segment with number m to point Р: The cross components E  and E  can be expressed by the projections , , xyz EEE:

Directional pattern, directivity factor and gain
The directional pattern (,) F   is determined by a field of the antenna in the radiation mode, the scattering pattern ( , ) s F   is determined by the field in the scattering mode. The fields in the radiation and scattering modes are described by the same expressions (30). Therefore, the directional pattern and the scattering pattern of the antenna are determined by the same expression: where max E is the maximum value of a field on a sphere Rc o n s t  .
The shape of (,) Numerical Simulation -From Theory to Industry 166 The antenna gain ( G ) characterises the antenna directional properties and power heat loss in antenna elements [Mittra, 1973]:

Radar Cross Section
The radar cross section (RCS) of antennas can exceed considerably the effective area of an object, on which the antennas are installed. That is why it is necessary to take into consideration the scattering properties of the antenna, when developping countermeasures to radar reconnaissance. In accordance with [Crispin and Maffett, 1965;Scolnic, 1970;] where S is the radar cross section; R is the distance from an antenna to a point of observation in space; E is the amplitude of a scattered field in the point of observation; i E is the amplitude of the electric field of an irradiating wave in the place of antenna location.
The scattering pattern and RCS depend on the antenna structure, frequency, polarization, and direction of falling of an electromagnetic wave on the antenna. RCS is described in general by the polarization scattering matrix: where S  is the RCS calculated according to a scattered field component E  , when the object is irradiated by a field i E  ; S  is the RCS calculated according to a scattered field component E  , when the object is irradiated by a field i E  ; S  is the RCS calculated according to a scattered field component E  , when the object is irradiated by a field i E  ; S  is the RCS calculated according to a scattered field component E  , when the object is irradiated by a field i E  .
The scattered field and RCS of the antenna can be represented as the sum of two components: antenna component Shkolnikov , 1974]. The irradiating wave creates conduction currents in the antenna conductors. These currents radiate the field that is a structural component of the scattered field. The RCS corresponding to this field is s S . At the same time, the antenna currents excite the transmission line connected to the antenna. The wave in the transmission line is reflected from a load and is radiated by the antenna. This scattered field is the antenna component. The RCS corresponding to this field is a S . If the load is matched to the transmission line, a S =0. The antenna components and the structural components of the scattered field differ from each other not only by their scattering pattern, but by their phase diagram as well. Therefore, after having determined separately the components a S , s S and the scattered fields corresponding to these components, it is impossible to found the total RCS as their arithmetical sum [Crispin and Maffett, 1965]. The antenna component of the RCS is related to an antenna gain and a VSWR in the transmission line. When calculating the RCS by the formula (34), it is necessary to connect the matched load to the antenna port. Such an oppotunity is provided in the integral equation (1). The number of the segment k , to which the load lk Z is connected, shall coincide with the number of a segment of excitation.
The scattering pattern differs from the directional pattern (DP) of an antenna in the transmission (reception) mode. It is related to the fact that a structural component predominates over an antenna component in a scattered field. The structural component of the scattered field is the result of radiation of the currents created by an incident EMW in the antenna. Distribution of these currents can differ essentially from current distribution in the antenna in the transmission mode.

Interaction of antennas located in near-field region
An integral equation and the subsequent part of a mathematical model of a wire antenna allow modelling wire antennas with an arbitrary geometry, located in the nearfield region. If several antennas are located in the near-field region, the length L is the total length of conductors of all the antennas. If, in the radiation mode, only one antenna from group functions (an active antenna), and entries of all the remaining antennas are loaded with matched loads (passive antennas), it is possible to determine an isolation ratio between the active antenna and every other passive antenna. The isolation ratio of a field is calculated as the ratio of complex amplitudes of currents at the entry of the active antenna () inp a I and a passive antenna () inp p I . The power isolation ratio p K is equal to: The described mathematical model has been used in the program, with the help of which the results below have been obtained. Fig.5 shows a Yagi-Uda antenna. Antenna elements: 1 is a linea reflector; 2 is a dipole; 3 are directors. Plane and corner reflectors are used too( see fig. 6). The polarization of the antenna is linear. Plane XZ is Е-plane; plane YZ is Н-plane. According to fig. 4, in XZ-plane  =0, in YZ-plane  =90°. The antenna is optimised as for its input resistance, gain, backscattered radiation level («Front to back» parameter: F/B). Its directional pattern, input resistance, directivity factor, gain depend on the antenna geometrical parametres, frequency, the surface resistance of conductors and the connected lumped resistances. Further, all the regularities are illustrated by the example of antennas intended for a medium frequency f=300 MHz. Size values of the antenna elements are not shown because of the lack of space. elements are selected so that the input resistance was equal to inp R  50 Ohm, inp X  0. The results have been obtained for the case s Z =0. Fig. 8 shows (as an example) a directional pattern in the E-plane at N=10. Fig. 9 shows the dependences of gain and «Front to Bback» ratio (F/B) on the reflector sizes in the Н-plane at a different distance between conductors of the reflector: h D =0,05 λ = 50 mm and h D =0,1 λ = 100 mm; fig. 10 shows the dependence of G and F/B on the reflector size in the E-plane.

Radiation characteristics
It follows from the cited and other results of simulation that the antenna gain and input resistance depend little on reflector sizes. If e R changes in the interval (0,2 … 1)λ , the active input resistance varies in the interval (48,5 … 50) Ohm, and the reactive input resistance varies in the interval (0 …-1,7) Ohm. If the distance between the conductors of the reflector is h D <0,1 λ, the antenna parametres hardly depend on Dh . The maximum of the F/B parametre corresponds to eh RR   (0,6 … 0,7) λ.  fig. 11 that when an angle  decreases, the F/B parametre worsens, and that the gain hardly depend on  . When  increases, the input resistance increases.

Scattering characteristics
By the example of a dipole, it is convenient to show the dependence of current distribution in the scattering mode on the direction of radiation. Fig. 12 shows current distribution in a dipole with nonresonant length L=1,25λ. The dipole is irradiated from the direction that is perpendicular to the dipole axis (along the Z axis, see fig. 5), and at an angle of 45° to the axis. Fig. 13 shows the corresponding scattering patterns in the E-plane.     fig. 15 that a structural component of RCS is much more bigger, than its antenna component.

Dipole antennas with reduced dimensions
In case of increase of a surface inductive reactance s X , resonance frequency fo of an antenna decreases. The most practical version of implementation of the surface resistance is the    , the reactance s X turns out to be equal to [Yurtsev, Runov, Kazarin 1974 The more the ratio / oo A H , the more s X and the less the antenna resonance frequency. Fig.  17 shows the dependence inp R and inp X on frequency at two values of o A for a dipole tuned in resonance at a frequency of 300 MHz at s X =0.
In case of decrease of Ho and Aoc o n s t  , the resonance frequency and a frequency band according to a matching criterion decreases. Fig. 18 shows the dependence of a resonance frequency o f for a Yagi-Uda antenna with a linear reflector and one director, depending on Ho at 0,003 3 o A   mm. The antenna of a smooth conductor with a radius o A =3 mm was tuned to a frequency of 300 MHz. In case of decrease of Ho , the resonance frequency o f decreases, the band properties worsen, the F/B parametre decreases, the antenna gain G decreases. Fig. 19 shows the dependence of the percentage bandwidth / o df f , in which  In case of change of o H from 10 mm to 2.5 mm, G decreases from 7 dB to 5 dB, F/B decreases from 20 dB to 14 dB.

Flat and convex dipole and Yagi arrays
One of the main questions of the numerical analysis of antenna arrays and antennas as a part of a group in the near-field region is the question of interaction of antennas. The  coupling effect of antennas, located in a near-field region, results in changes of all the characteristics of the antennas. Further, the differences in interaction between emitters in plane and convex arrays have been studied by the example of comparison of linear and arc arrays. The interaction of two Yagi-Uda antennas has also been studied, depending on a distance between them and their cross orientation. It is important for the organisation of communication nodes.
The received regularities are illustrated by the example of a Yagi-Uda antenna with the number of directors N=1. The antenna is tuned so that the input resistance is equal to 50 Ohm at a medium frequency of 300 MHz. The numerical simulation shows that, with the increase of the number of the emitters in the fragment, the shape of the directional pattern in some angular sector tends to become rightangled. The degree of interaction of two emitters between themselves can be evaluated by the isolation ratio 2 12 12 ( / ) SI I  , where 1 I is the amplitude of an input current of an emitter in the radiation mode, 2 I is the amplitude of an input current of a passive emitter loaded with a matched load. Fig. 22 shows the dependence 12 S on '/ RR   , where R is the distance between emitters. The calculations have been made for a Yagi-Uda antenna at the number of directors N=1 and N=5 with the location of antennas in the E-plane and the H-plane. It follows from the simulation results that 12 S depends little on the number of directors. If the antennas are located in the H-plane, the interaction is stronger. If the distance is R   , the interaction of the antennas is negligible. It means that it is sufficient to take into consideration the interaction of a Yagi-Uda antenna as a part of a plane array only with an environment of two rings. The influence of interaction on the directional pattern of a Yagi-Uda antenna is different in plane and convex arrays. Fig. 23 shows the directional patterns in the E-plane of a director antenna with N=1 in the arc array for the number of emitters М=3 and М=5 at a distance between emitters D=0,55  . The angle between the axes of the neighboring antennas is equal to β =22°. In both cases, the middle antenna is excited, the remaining ones are loaded with matched loads.
It follows from the comparison of fig. 20b, c and fig. 23b, c that, at the same linear distance between the neighboring emitters (D), the interaction between director antennas as a part of an arc array results in more considerable changes, than if they are a part of a linear array. The dependence of an isolation ratio between two director antennas on an angle between their axes of antennas (β) is shown in fig. 24 for two distances between the antennas entries D' =D/  = 0,5 and D' =D/  = 0,7 and two values of the number of directors N=1 and N=5. , when an angle β increases, the isolation ratio decreases at first, then it increases and reaches a maximum at β =110°-120°. 2. If (0,65 0,7) D   , when an angle β increases, the isolation ratio increases at first, then it decreases and reaches a minimum at β  90°.

Linear Antenna Array with Series Excitation
The dipole arrays of half-wave dipoles with series excitation are decribed briefly in the book [Rothammels, 1995] with the reference to the works [Brown, Liwis and Epstein, 1937;Bruckmann, 1938). Such antennas are gone by the name of Franklin antennas (Franckin C.S. -British patent № 242342, 1924). Versions of such antennas are shown in fig. 25. The drawing symbols are the following: A is half-wave dipoles: B is closed quarter-wave stubs; C is double-wire line exciting line; D is stub bonding; L is dipole length; H is stub length; d is stub width; P is point of observation in space; P1 is projection of point Р on XY plane; R, θ, φ are the spherical coordinates of point P.  The polarization of an antenna radiation field is linear: the E-plane is the plane φ=const, the H-plane is the plane XY. In the H-plane, the directional pattern is quasi-isotropic.
Antenna characteristics depend on geometrical sizes of L, H, d, the number of dipoles in the antenna, an angle between the neighbouring stubs, frequency f. In the literature, these regularities are not studied, and they are the subject of the numerical simulation in the present chapter. Besides, in the chapter, we study the modifications of a Franklin antenna that are more wideband as for the matching criterion and have a sector-shaped directional pattern in the E-plane.
The version shown in fig. 25a at a resonance frequency, which corresponds to the equality to zero of a reactive part of an input resistance (Х=0), has a high active input resistance (R f =1000 MHz) and two cases with the number of dipoles N=3 and N=5. It is evident that at N=3, there is a frequency, at which Х=0. At N  5, there is no such a frequency (Х <0). It complicates the antenna matching with a transmission line.
The more convenient version for matching is the one, in which the exciting voltage is introduced into a bonding rupture. Such a version is shown conditionally in fig. 27. The place of the antenna excitation is marked by a big point. The dependences R(f) and X(f) are shown in fig. 28 a. In such an antenna, the choice of values L, H, d can be X=0, and R=50 Ohm or 75 Ohm. The graphs of fig. 28 correspond to the antenna tuning to a frequency of 1000 MHz in order to obtain R=50 Ohm, X=0 (L=142 mm; H=71,5 mm; d=8 mm). The symbols of R and X are the same as in fig. 26.
The numerical simulation shows that with increase of the number of dipoles in the antenna (N), the antenna band properties worsen. The VSWR minimum does not match to the most acceptable directional pattern in the E-plane. Fig. 29 shows the directional patterns at three frequencies of the band, in which VSWR<2. The frequency values, gain (G), R, X, VSWR are also given. The graphs of directional patterns in the H-plane give the values of nonuniformity of the directional pattern (Nu). The nonuniformity Nu decreases, if to direct the neighbouring stubs in the contrary directions ( fig. 25b). Antenna characteristics depend on a number of an excited stub. In fig. 27, the central stub is excited. If to displace an excitation point from the central stub, the maximum of a directional pattern in the E-plane declines from a normal to the Z axis. Fig. 29 illustrates these properties by the example of the antenna with N=6. When the frequency increases, the angle of deflection of the maximum of a directional pattern decreases. Fig. 30 shows the DP of the antenna with N=13 within a bandwidth, in which VSWR<2. The lower stub is excited, as in fig. 29. The band of matching frequencies depends little on the number of dipoles in the antenna, if the extreme stub is excited. For illustration, fig. 31 shows the dependence of VSVR(f) at different values of the number of dipoles.
When the number of dipoles increases, the misphasing between the stub that is excited and the extreme stubs increases. Therefore, when the number of dipoles increases, the antenna gain at first increases promptly, then slowly. The dependence of gain (G) on the number of dipoles is shown in fig. 32. At N=1, the antenna represents a half-wave dipole that has G=2 dB. The lower stub is excited in the antenna. It follows from fig. 32 that there is no sense to make the number of dipoles N> 5.  If to apply a system of reflectors or (and) directors to an antenna, it is possible to make a directional pattern sector-shaped in the H-plane. Fig. 33 shows such an antenna and its directional pattern.
A Franklin antenna and its modifications can be used as transmitting antennas in radio links. In the literature, the methods of antenna excitation were studied, their key properties were studied briefly [Rothammels K, 1995], but there is no information about the main regularities. Further, these regularities are examined for version 34а only, because this version is generally used.  It follows from fig. 36а that, by selecting the values of L and β at preset Н, it is possible to make input resistance R preset and to meet the condition 0 X  .

Loop antenna of closed and open arrays
The antenna band properties are illustrated by fig. 37, 38. The calculation has been made for the antennas calculated so that at a frequency of 300 MHz the input resistance was equal to R  50 Ohm, 0 X  , see fig. 37a, and R  75 Ohm, 0 X  , see fig. 37b. Fig. 38a shows the dependence of VSWR on frequency. For antenna 1 with the input resistance of 50 Ohm at a frequency of MHz, VSWR has been calculated in a line with the characteristic resistance of 50 Ohm (graph 1), for antenna 2 with the input resistance of 75 Ohm, VSWR has been calculated in a line with the characteristic resistance of 75 Ohm (graph 2). It is evident that the antenna with a higher input resistance is matched to a bigger bandwidth. Fig. 38b shows the dependences of gain and F/B parametre for the second antenna.

Interaction of loop antennas as part of antenna array
Interaction of loop of antennas as a part of an array results in changing of their characteristics. It shall be taken into consideration, when designing antenna arrays. Fig. 39 shows directional patterns in the Е and Н-planes of a loop antenna with an input resistance of 50 Ohm: isolated one (a); when taking into account its interaction with two neighbouring emitters -one on the left and one on the right (b); and taking into account its interaction with four neighbouring emitters -two on the left and two on the right. The values of input resistance, gain and parametre F/B are also shown in the figures. The calculations have been made for a linear array, in which emitters are located in the E-plane, at a frequency of 300 MHz (as an example). The distance between the neighbouring emitters is equal to 0,65λ. It follows from the results of simulation that the interaction influences an input resistance and F/B parametre insignificantly. The directional pattern and gain influence the interaction greatly, but differences in influence of various number of interacting loop antennas are insignificant ( fig.39b and 39c). It means that when simulating multiple-unit arrays of loop antennas, it is possible to take into consideration interaction of each emitter only with its nearest emitters, and further to use the theorem of multiplication of directional patterns. The linear distance between the neighbouring emitters is equal to 0.65λ, the angle between axes of antennas is equal to 12°. It follows from the comparison of fig. 39 and fig. 40 that interaction between loop antennas in an annular array is less than interaction in a linear array.

Linear antenna arrays with coherent excitation
To increase gain, it is possible to use linear arrays of loop antennas with series excitation. . Fig. 42b illustrates the dependence of the input resistance on the number of arrays. The sizes of elements of the arrays are chosen so that the input resistance was equal to 50 Ohm at Мp=2. It follows from the results of simulation that it is not reasonable to make Mр>4 in the antenna. For each M value, it is necessary to optimise the sizes of elements of arrays in order to obtain the preset input resistance.
In a linear array consisting of arrays with a lateral length 0,5 L   , the gain is 3 dB more approximately. Fig. 43а shows the version of a linear array with a quasi-isotropic directional pattern in the E-plane. Such a pattern can be obtained, if to bend all the arrays along the Z axis at an angle  =35-40°. It is possible to ensure the preset input resistance by choosing the sizes of elements. Fig. 43b,c shows the directional patterns and the values of the parameters of G (Gain), NR (nonuniformity of a directional pattern in the E-plane), R, X, values of VSWR in a line with a wave resistance of 50 Ohm.

Conclusion
The simulation results, given in this chapter, complete the information about dipole and loop antennas that is available in the literature, and can be used to choice a type of an antenna according to specified requirements and after estimating its main characteristics. Numerical simulation of an antenna can be done with the use of the formulas given in the description of a mathematical model. When using a program developed on the basis of these formulas, it is necessary to carry out research into the convergence of the results of calculation. The most sensitive parametre of the antenna in relation to the number of segments of division of conductors (M) is the input resistance. When using impulse functions as the basis and weight ones, it is possible to be oriented towards the following recommendations: the ratio of a segment length to a conductor diameter / 2 LA o   0,8-1,2; the number of segments at a wavelength: 80-200.
In some cases, the results given in the graphs can be used directly with the use of electrodynamic scaling.
We express aur gratitude to D.Moskalev and V.Kizimenko for the useful discussion of the results described in this chapter.