Driving Control Technologies of New High-Efficient Motors

Abstract

Although switched-reluctance machine (SRM) possesses many structural advantages and application potential, it is rather difficult to successfully control with high performance being comparable to other machines. Many critical affairs must be properly treated to obtain the improved operating characteristics. This chapter presents the basic and key technologies of switched-reluctance machine in motor and generator operations. The contents in this chapter include: (1) structures and governing equations of SRM; (2) some commonly used SRM converters; (3) estimation of key parameters and performance evaluation of SRM drive; (4) commutation scheme, current control scheme, and speed control scheme of SRM drive; (5) some commonly used front-end converters and their operation controls for SRM drive; (6) reversible and regenerative braking operation controls for SRM drive; (7) some tuning issues for SRM drive; (8) operation control and some tuning issues of switched-reluctance generators; and (9) experimental application exploration for SRM systems: (a) wind generator and microgrid; (b) EV SRM drive.

1. Introduction

Induction motors [1, 2, 3] are still the most popularly applied motor owing to their simple structures, ease of operation, and mature driving technologies. However, their best efficiencies are generally still in IE3-class due to the existence of rotor copper losses. Upgrading the efficiencies of IMs to IE4-class via using high-grade core and rotor cage materials will not be cost-effective. Permanent-magnet synchronous motor (PMSM) and synchronous reluctance motor (SynRM) [2, 3, 4, 5, 6] are the two new motors being able to achieve the IE4-class and even the IE5-class efficiency. As to the switched reluctance motor (SRM) [7], although it may also possess the potential, it still lacks commercialized motors and power modules being available.

To establish a high-performance PMSM drive, the proper match between the available source, the interface converter, the motor drive, and the mechanical load must be made. Some major affairs to be properly treated include: (i) physical modeling and parameter estimation [8, 9]; (ii) current control [10, 11, 12, 13]: generally, the current control methods can be categorized into hysteresis controls, ramp-comparison controls and predictive controls; (iii) speed control [14, 15, 16, 17]; (iv) direct torque control based on space-vector PWM [18]; (v) commutation shift and field-weakening control [19, 20, 21, 22, 23, 24, 25]: through appropriate d-axis field current setting or commutation shift, the developed torque can be increased by utilizing the reluctance torque component effectively. In the case of constant power, it can extend the speed range through field-weakening; (vi) voltage boosting [26, 27, 28]: for a battery-powered EV PMSM drive, the equipped battery followed interface DC/DC converter providing adjustable DC-link voltage can increase the battery voltage selection flexibility and the overall drive system rating utilization. The buck-boost bidirectional interface converters are adopted in the PMSM motor drive [27]. It can step down the voltage from battery; hence the lower DC-link voltage under lower speed can improve the efficiency. As the motor drive is drawn from the mains, the single-phase or three-phase switch-mode rectifiers (SMRs) [29, 30, 31, 32] must be equipped to achieve the DC-link voltage boosting task.

For a standard PMSM drive, the rotor absolute position is necessary to make its vector control. However, for avoiding the risk of sensor failure, the position sensorless controlled motor drive is preferable. Generally, the existing PMSM position sensorless control approaches can be classified into [33, 34, 35]: (i) methods based on the measured and/or identified machine parameters [36, 37]; (ii) methods based on rotor saliency and magnet anisotropy: the high-frequency injection approaches are the typical ones. The high-frequency sinusoidal-wave [38, 39, 40] or square-wave [41, 42] voltage with suited amplitude and frequency is injected into the studied motor, and the detected current is processed to yield the observed rotor position. These approaches can be successfully operated under low speed including standstill, and no machine parameters are needed. However, the injected signal will yield high-frequency noise, lower efficiency and the back-EMF harmonics in IPMSM may cause the estimated rotor position error; (iii) observer based methods [36, 43, 44]: these include adaptive observers, sliding mode observers, Kalman filter observers and reduced-order observers, etc.; (iv) back-EMF methods [45, 46, 47]: these methods are obviously not suitable for EV due to its frequent accelerating/decelerating operations; and (v) hybrid methods [48, 49].

As generally recognized, PMSM possesses high power density and efficiency. However, its manufacture process is complicated with higher cost. And it has the demagnetization risk. As to the synchronous reluctance motor (SynRM), although it has no permanent magnet torque component, good performance/cost-compromised performance can still be obtained if the proper switching control is made. It possesses high application potential, especially for high-speed driving, since its rotor is rigid and cogging torque free from having no permanent magnets and conductors.

Since only some air slots are existed in the rotor of a SynRM, its energy conversion characteristics are significantly affected by rotor geometry [50, 51, 52]. The rotor is assembled with stacked silicon steels, which leads the rotor to be easily saturated, and the iron loss is comparatively higher than PMSM. Hence, considering iron loss and magnetic saturation is necessary for achieving high performance and efficiency.

In efficiency optimization of a SynRM, both the copper loss and the iron loss must be considered. The copper loss is load-dependent, while the iron loss is related to the air-gap flux level. Basically, loss minimization approaches [53, 54, 55, 56, 57, 58, 59, 60, 61, 62] for SynRM can be roughly categorized into loss model-based control and searching control. The former approaches minimize the power consumption by setting the optimal d-axis current command, which is derived from the loss function using machine parameters. Obviously, the resulted control performance is affected by the motor parameter variations, especially the d-axis inductance. As to the searching methods, their performances are insensitive to motor parameter variations. However, the complicated searching process usually causes the sluggish dynamic response, torque ripple, and additional losses. Finally, for illustrating the key technological affairs introduced in this article, the example battery/supercapacitor powered EV PMSM drive [63] and SMR-fed SynRM drive [64, 65] are presented in Section 5 and Section 6.

2. Basic motor drive system configuration

As shown in Figure 1, motor drive is an interdisciplinary mechatronic system including motor, mechanical load, power converter, control scheme, transducing, and sensing schemes. The proper design of motor and the proper match between system-constituted components should be made for yielding good driving performance.

Figure 1.

Typical motor drive system configuration.

The commonly used motors include brushed DC motor (DCM), induction motor (IM), and synchronous motor (SM) consisting of permanent-magnet synchronous motor (PMSM), synchronous reluctance motor (SynRM), switched reluctance motor (SRM), etc. The ideal developed torque characteristics of some typical motors are sketched in Figure 2. In reality, each type of motor possesses its distinct affairs, which must be properly treated.

Figure 2.

Sketched ideal developed torque characteristics of some motors: (a) DCM, (b) SRM, (c) three-phase sine-wave PMSM (d) three-phase square-wave PMSM.

3. Permanent-magnet synchronous motors

3.1 Structural features of synchronous motors

Except for the slot-less stator, the slotted and salient-pole stators are shown in Figure 3a and b. The former is mainly employed for forming distributed armature winding with sinusoidal currents. It is suitable for low-speed driving applications because it generates smoother developed torque. The salient-pole stator in Figure 3b is used for concentrated windings. This winding has higher efficiency owing to its lower copper losses. However, it possesses higher torque ripple and thus only suited for speed driving applications.

Figure 3.

Typical stator structures of synchronous motors: (a) slotted and (b) salient-pole.

Some rotor structures of SM are sketched in Figure 4a–g. The SPMSM shown in Figure 4a possesses less cogging torque and smoother developed torque. Hence, it is suitable for low-speed driving applications. In order to increase the rigidity, one can adopt the inset SPMSM shown in Figure 4b. There are many types of interior PMSM (IPMSM) with various shapes of PM. The two typical rotor structures of IPMSM with buried and interior magnets are shown in Figure 4c and d. The existence of saliency allows them to produce reluctance torque in addition to electromagnetic torque. And the field weakening is simpler to achieve. For higher speed driving applications, one can adopt the permanent-magnet-assisted SynRM (PMa-SynRM) shown in Figure 4e. Two types of SynRM rotor structure are shown in Figure 4f and g, namely, the axially laminated anisotropy (ALA) rotor and the transversally laminated anisotropy (TLA) rotor.

Figure 4.

Some typical rotor structures of synchronous motors: (a) SPMSM, (b) inset SPMSM, (c) radial IPMSM, (d) tangential IPMSM, (e) hybrid V-shape IPMSM, (f) PMASynRM, (g) ALA SynRM, and (h) TLA SynRM.

Comments: SynRM belongs to SM having: (i) distributed armature windings; and (ii) PM-free rotor with air slots. As to the switched reluctance motor (SRM), its features lie in: (i) concentrated armature windings and (ii) teethed rotor. It follows that these two reluctance machines have the major comparative features: (i) they all have only reluctance torque, and the developed torque or power is highly affected by commutation instant setting and (ii) although they all possess high ripple torques under low speed, the one yielded by SynRM is less owing to distributed armature with sinusoidal current excitation.

3.2 Physical modeling of PMSM

3.2.1 Voltage equations

Figure 5 shows the configuration of a Y-connected three-phase 2-pole PMSM. The Y-connected resistors across the armature terminals are used to estimate the motor phase back-EMF under no-load, that is, $v_{a{n}^{\prime }}\approx e_{\mathit{as}}$. The following assumptions are made for making analytic derivation: (i) symmetrical and sinusoidally distributed three-phase armature windings, (ii) sinusoidal armature back-EMFs, and (iii) linear magnet circuit. And the rotor position ${\theta }_r$ is defined as the angle between as-axis and the sensed q-axis.

Figure 5.

Configuration of an elementary three-phase two-pole Y-connected PMSM.

The phase (a-phase as an example, the ones for the other two phases, can be written analogously) voltage equation can be expressed as:

$$v_{\mathit{as}}=R_s{i}_{\mathit{as}}+\frac{d}{\mathit{dt}}{\lambda }_{\mathit{as}}=R_s{i}_{\mathit{as}}+\frac{d}{\mathit{dt}}\left(L_{\mathit{\text{asas}}}{i}_{\mathit{as}}+L_{\mathit{\text{asbs}}}{i}_{\mathit{bs}}+L_{\mathit{\text{ascs}}}{i}_{\mathit{cs}}\right)+e_{\mathit{as}}$$E1

$$L_{\mathit{\text{asas}}}=L_{\mathit{ls}}+L_A+L_{B}cos2{\theta }_r$$E2

$$e_{\mathit{as}}={\lambda }_m^{\text{'}r}{\omega }_{r}cos{\theta }_r\underset{¯}{\underset{¯}{\mathrm{\Delta }}}{\widehat{E}}_{\mathit{as}}cos{\theta }_r$$E3

$${\theta }_r={\theta }_r\left(0\right)+{\int }_0^t{\omega }_r\left(\zeta \right)\mathit{d\zeta }$$E4

where $v_{\mathit{as}}$ is winding terminal voltages, $i_{\mathit{as}}$ is winding current, $R_s$ is winding resistance, ${\lambda }_{\mathit{as}}$ is winding flux linkage, ${\lambda }_m^{\text{'}r}$ is peak flux linkage contributed by the rotor permanent magnet, $L_{\mathit{\text{asas}}}$ is winding self-inductance, $L_{\mathit{\text{asbs}}}$ is mutual inductance, $L_{\mathit{ls}}$ is winding leakage inductance, ${\theta }_r$ is rotor electrical angular position, ${\theta }_r\left(0\right)$ is the initial position of q-axis relative to a-axis, ${\omega }_r$ is rotor electrical angular speed, $e_{\mathit{as}}$ is back electromotive force (EMF), and ${\widehat{E}}_{\mathit{as}}$ is the peak value of $e_{\mathit{as}}$.

By applying rotor rotating frame transformation, the voltage equations in dq-frame can be written as:

$$v_{\mathit{qs}}=\left(R_s+{\mathit{pL}}_q\right)i_{\mathit{qs}}+{\omega }_r{L}_d{i}_{\mathit{ds}}+{\omega }_r{\lambda }_m^{\text{'}r}$$E5

$$v_{\mathit{ds}}=\left(R_s+{\mathit{pL}}_d\right)i_{\mathit{ds}}-{\omega }_r{L}_q{i}_{\mathit{qs}}$$E6

$$L_q=L_{\mathit{ls}}+L_{\mathit{mq}}=L_{\mathit{ls}}+\frac{3}{2}\left(L_A+L_B\right)$$E7

$$L_d=L_{\mathit{ls}}+L_{\mathit{md}}=L_{\mathit{ls}}+\frac{3}{2}\left(L_A-L_B\right)$$E8

where $L_q\left(L_d\right)$ denotes q-axis (d-axis) inductance and $L_q>L_d$.

3.2.2 Torque and mechanical equations

The electromagnetic developed torque and mechanical equations of a PMSM drive in dq-frame can be expressed as:

$$T_e=\frac{3P}{4}\left[{\lambda }_m^{\text{'}r}{i}_{\mathit{qs}}+\left(L_d-L_q\right)i_{\mathit{qs}}{i}_{\mathit{ds}}\right]=\frac{3P}{4}\left[{\lambda }_m^{\text{'}r}{\widehat{I}}_{\mathit{as}}cos\beta +\frac{L_d-L_q}{2}{\widehat{I}}_{\mathit{as}}^{2}sin2\beta \right]=T_L+B{\omega }_r+J\frac{d{\omega }_r}{\mathit{dt}}$$E9

where $P$ is pole number, ${\widehat{I}}_{\mathit{as}}$ is peak of a-phase current, the variables $\beta$ denotes the shift angle between q-axis and peak of a-phase current, $T_L$ is load torque, $B$ is total damping coefficient, and $J$ is total inertia constant.

3.3 Measurement of motor key parameters

Before establishing the motor drive, some key motor parameters must be measured to comprehend the characteristics of the studied motor. Figure 6 shows the experimental mechanism for conducting the measurements.

Figure 6.

Parameter measurement mechanism for SMs.

3.3.1 Winding resistance and inductances

For the Y-connected PMSM with isolated neutral as indicated in Figures 5 and 6, from Eq. (1), the inductance and resistance of PMSM between a-phase and b-phase can be expressed as:

$$R_{\mathit{ab}}=2R_s,L_{\mathit{ab}}\left({\theta }_r\right)=2L_{\mathit{ls}}+3L_A-3L_{B}cos\left(2{\theta }_r-\frac{2\pi }{3}\right)$$E10

And the q-axis and d-axis inductances of Eqs. (7) and (8) can be expressed from Eq. (10) as:

$$L_q=\frac{L_{\mathit{\text{asbs}},max}}{2}=\frac{1}{2}{L}_{\mathit{\text{asbs}}}\left({\theta }_r=-\frac{\pi }{6}\right)=L_{\mathit{ls}}+\frac{3}{2}{L}_A+\frac{3}{2}{L}_B$$E11

$$L_d=\frac{L_{\mathit{\text{asbs}},min}}{2}=\frac{1}{2}{L}_{\mathit{\text{asbs}}}\left({\theta }_r=\frac{\pi }{3}\right)=L_{\mathit{ls}}+\frac{3}{2}{L}_A-\frac{3}{2}{L}_B$$E12

1. Winding resistance: the armature winding resistance $R_s=R_{\mathit{ab}}/2$ can be estimated by DC excitation method.

2. Winding inductances: the $L_{\mathit{ab}}\left({\theta }_r\right)$ can be measured using the LCR meter under various frequencies at different rotor positions. Then the q-axis and d-axis inductances $L_q$ and $L_d$ can be obtained from Eqs. (11) and (12). However, the inductance-saturated effects cannot be acquired for the small-signal excitation. On the other hand, one can apply the step-response method under various current levels.

3.3.2 Back-EMF and PM flux linkage ${\lambda }_m^{\text{'}}$

As shown in Figure 5, three Y-connected resistors with a reasonably high resistance (R = $100\mathrm{k\Omega }$) are connected at the stator terminals to observe the motor phase back-EMF under no-load. By letting $i_{\mathit{as}}=i_{\mathit{bs}}=i_{\mathit{cs}}=0$, the back-EMF can be found from Eqs. (1) and (3) as:

$$e_{\mathit{as}}={\widehat{E}}_{\mathit{as}}cos{\theta }_r={\left.v_{\mathit{as}}\right|}_{\mathrm{NL}}\approx {\left.v_{a{n}^{\prime }}\right|}_{\mathrm{NL}}$$E13

And ${\lambda }_m^{\text{'}r}$ can be obtained from Eq. (3) as:

$${\lambda }_m^{\text{'}r}=\frac{{\widehat{E}}_{\mathit{as}}}{P/2\times {\omega }_r}\left(\mathrm{Wb}\right)$$E14

3.3.2.1 Measured examples

1. Figure 7a and b depict the measured back-EMFs and their spectra of two examples of PMSMs: (a) under 1000 rpm of a three-phase SPMSM, 5 kW, 24.5 N-m, 8-pole, 2000 rpm and (b) under 1000 rpm of a three-phase IPMSM, 1 kW, 3.23 N-m, 6-pole, 3000 rpm.

2. Observations: for the IPMSM, the distorted back-EMF waveform with eleventh and thirteenth orders are observed. The influences of back-EMF harmonics on the current control and the high-frequency-injected position sensorless control must be considered.

Figure 7.

Measured back-EMFs and their spectra of two examples of PMSMs: (a) under 1000 rpm of a three-phase SPMSM, 5 kW, 24.5 N m, 8-pole, 2000 rpm and (b) under 1000 rpm of a three-phase IPMSM, 1 kW, 3.23 N m, 6-pole, 3000 rpm.

3.4 Some key issues

Figure 8 shows some key issues affecting the performance of PMSM and SynRM drives. The typical ones include: (i) suitable motor type selection, (ii) motor parameters and dynamic model estimation, (iii) commutation instant setting and shifting, (iv) inverter and its PWM switching control, (v) field weakening, (vi) DC-link voltage boosting, (vii) regenerative braking operation, (viii) generator operation, and (ix) position sensorless control. Some comments are given below:

1. Motor selection: the stator and rotor structural features of some SMs have been explored in Section 3.1. The comparative loss and efficiency characteristics of some PMSMs and SynRM are depicted in Figure 9 [2, 3]. Basically, as the speed is increased, the PMSM with fewer permanent magnets is chosen for the ease of conducting field-weakening control. And finally, the SynRM or SRM is employed for higher speed driving applications.

2. Commutation: from Eqs. (3)–(9), one can be aware that the back-EMF and the developed torque of PMSM are highly affected by the speed, load, and commutation angle. The commutation shifting angle $\beta$ must be properly set and tuned to yield better developed torque. Commutation-advanced shift possesses field-weakening effects to reduce the back-EMF effects on the winding current response. It is worth noting that the commutation shift can also be equivalently achieved via directly setting the d-axis field current command. Basically, the commutation advanced shifting angles for various SMs are (i) SPMSM, $\beta =0^{\circ }$; (ii) SynRM, $\beta ={45}^{\circ }$ theoretically; and (iii) IPMSM and PMASynRM, $\beta =0\sim {45}^{\circ }$. This fact can also be observed from Figure 9.

3. Current control and effects of current ripple: for an inverter-fed AC motor drive, its driving characteristics are greatly affected by the adopted PWM approach, including fundamental frequency control and harmonic spectral features. The current-controlled PWM (CCPWM) scheme is normally applied for high-performance motor drive. As indicated in Figure 10a and b, the CCPWM scheme can be realized in abc-domain or dq-domain.

4. In some special application cases, the SPMSM is still adopted as the actuator for very high-speed driving applications with very compact volume. In these cases, the accurate commutation instant setting is very critical for avoiding magnet demagnetization, especially for position sensorless controlled drive. Moreover, the suited filtering chokes must be inserted between the inverter and the motor, as indicated in Figure 11.

5. Voltage boosting: under high speed, the sufficiently high back-EMF may make the winding current tracking performance sluggish as indicated in Figure 11 and thus yield the worsened developed torque. Voltage boosting is the effective means [26, 27, 28] to solve these problems. To accomplish this goal, one must equip the suited SMR or DC-DC interface converter for the motor drive being powered from the three-phase/single-phase mains or the battery. Some typical schematic arrangements of the bilateral SMRs and DC-DC interface converters are shown in Figures 12 and 13.

6. Generator operation of PMSM: according to Eq. (9), the generating mode of a PMSM is achieved as: PMSG: Q-axis torque current command is directly set to be negative; Regenerative braking and reversible operation of PMSM: the q-axis torque current is automatically set to be negative by the control scheme to achieve normal operations.

7. Sensorless control: basically, the most commonly used position sensorless methods are the observer back-EMF based and the HFI approaches. The former methods can only possess satisfactory running characteristics above a certain speed. Moreover, the motor must be started under traditional synchronous motor mode. By contrast, the HFI approach can be operated effectively in standstill and low speed, so this approach is applicable to frequent-starting occasions, for example, electric vehicles, tractions, elevators, etc. No motor parameters and externally added schematics are needed in constructing a HFI sensorless controlled SynRM drive. However, as mentioned above, the injected signal will yield acoustic noise and lower efficiency, and the back-EMF harmonics of IPMSM and SynRM may cause the estimated rotor position error.

Figure 8.

Some key issues of PMSM and SynRM drives.

Figure 9.

Comparative efficiencies of some typical synchronous motors.

Figure 10.

Configurations of CCPWM schemes in (a) abc-domain and (b) dq-domain.

Figure 11.

Effects of DC-link voltage level and non-ideal winding current waveforms on the AC motor drive driving characteristics.

Figure 12.

Motor drive with bidirectional SMR front end: (a) three-phase SMR and (b) single-phase SMR.

Figure 13.

Motor drive with bidirectional DC-DC converter front end: (a) one-leg boost-buck converter and (b) H-bridge converter.

4. Synchronous reluctance motors

4.1 Structural features

As shown in Figures 3 and 4, SynRM possesses distributed armature windings and PM-free rotor with air slots. The armature is exited with sinusoidal currents. Since it has only the reluctance torque, the developed torque is highly affected by commutation instant setting.

4.2 Physical modeling

4.2.1 Neglecting core loss

Different from the PMSM shown in Figure 5, the d-axis is conventionally chosen as the reference as indicated in Figure 14a. Following the similar deriving procedure described in the previous section, one can yield the voltage equations of SynRM:

Figure 14.

A two-pole three-phase synchronous reluctance motor: (a) configuration and (b) Y-connected armature circuit.

$$v_{\mathit{qs}}=\left(R_s+L_q\frac{d}{\mathit{dt}}\right)i_{\mathit{qs}}+{\omega }_r{L}_d{i}_{\mathit{ds}},v_{\mathit{ds}}=\left(R_s+L_d\frac{d}{\mathit{dt}}\right)i_{\mathit{ds}}-{\omega }_r{L}_q{i}_{\mathit{qs}}$$E15

$$L_q=L_{\mathit{ls}}+L_{\mathit{mq}}=L_{\mathit{ls}}+\frac{3}{2}\left(L_A-L_B\right),L_d=L_{\mathit{ls}}+L_{\mathit{md}}=L_{\mathit{ls}}+\frac{3}{2}\left(L_A+L_B\right)$$E16

with $L_{\mathit{mq}}$ ($L_{\mathit{md}}$) being the q-axis (d-axis) magnetizing inductances, respectively. Obviously, $L_d>L_q$ can be observed from Eq. (16).

The developed torque and mechanical equations of a SynRM drive are:

$$T_e=\frac{3P}{4}\left(L_d-L_q\right)i_{\mathit{qs}}{i}_{\mathit{ds}}=\frac{3P}{4}\frac{L_d-L_q}{2}{\widehat{I}}_{\mathit{as}}^{2}sin2\beta =T_L+B{\omega }_r+J\frac{d{\omega }_r}{\mathit{dt}}$$E17

where ${\widehat{I}}_{\mathit{as}}$ is peak of a-phase current, the variable $\beta$ denotes the shift angle between the d-axis and the peak of a-phase current, $i_{\mathit{qs}}={\widehat{I}}_{\mathit{as}}sin\beta$ and $i_{\mathit{ds}}={\widehat{I}}_{\mathit{as}}cos\beta$.

Comments: from Eq. (17) one can find some facts: (i) it only possesses the reluctance torque, which is nonlinear and (ii) although $\beta =45°$ can be set to yield the maximum torque, theoretically, the proper setting of $\beta$ is required for maximizing $T_e$ in reality.

4.2.2 Considering core loss

For a SynRM, the core loss is more significant than other motors. The d-axis and q-axis equivalent circuits considering iron loss with $R_c$ are shown in Figure 15a. And Figure 15b shows the phasors of torque currents and terminal currents considering iron loss. From which, the terminal currents $i_{\mathit{qs}}$ and $i_{\mathit{ds}}$ can be expressed using the torque currents $i_{\mathit{mq}}$ and $i_{\mathit{md}}$ as:

Figure 15.

Basic characteristics of SynRM: (a) equivalent circuits considering iron loss and (b) phasors of voltages and currents neglecting core loss.

$$\left[\begin{array}{c}{i}_{\mathit{qs}}\\ i_{\mathit{ds}}\end{array}\right]=\frac{L_d{L}_q}{R_c}\left[\begin{array}{cc}\left(s+R_c/L_q\right)/L_d& {\omega }_e/L_q\\ -{\omega }_e/L_d& \left(s+R_c/L_d\right)/L_q\end{array}\right]\left[\begin{array}{c}{i}_{\mathit{mq}}\\ i_{\mathit{md}}\end{array}\right]$$E18

And the steady-state expressions of Eq. (4) are:

$$\left[\begin{array}{c}{I}_{\mathit{qs}}\\ I_{\mathit{ds}}\end{array}\right]=\left[\begin{array}{cc}1& L_d{\omega }_e/R_c\\ -L_q{\omega }_e/R_c& 1\end{array}\right]\left[\begin{array}{c}{I}_{\mathit{mq}}\\ I_{\mathit{md}}\end{array}\right]$$E19

The developed torque of a SynRM using $i_{\mathit{mq}}$ and $i_{\mathit{md}}$ can be expressed as:

$$T_e=\frac{3P}{4}\left(L_d-L_q\right)i_{\mathit{mq}}{i}_{\mathit{md}}$$E20

4.2.2.1 Motor losses

From Figure 15b, the copper loss $P_{\mathit{cu}}$, the core loss $P_c$, and the total loss $P_t$ of a SynRM can be expressed as:

$$P_{\mathit{cu}}=\frac{3}{2}{R}_s\left(i_{\mathit{qs}}^2+i_{\mathit{ds}}^2\right),P_c=\frac{3}{2}{R}_c\left(i_{\mathit{cq}}^2+i_{\mathit{cd}}^2\right)$$E21

$$\begin{array}{l}{P}_t=P_{\mathit{cu}}+P_c=\frac{3}{2}\{\left[R_s+\frac{{\left({\omega }_e{L}_q\right)}^2}{R_c}+\frac{{\left({\omega }_e{L}_q\right)}^2{R}_s}{R_c^2}\right]i_{\mathit{mq}}^2+\left[R_s+\frac{{\left({\omega }_e{L}_d\right)}^2}{R_c}+\frac{{\left({\omega }_e{L}_d\right)}^2{R}_s}{R_c^2}\right]i_{\mathit{md}}^2\\ +\left[\frac{2R_s}{R_c}{\omega }_e\left(L_d-L_q\right)\right]i_{\mathit{mq}}{i}_{\mathit{md}}\}\end{array}$$E22

By defining the variable $\varsigma =i_{\mathit{mq}}/i_{\mathit{md}}$, the commutation angle $\beta ={\beta }_o$ is derived to yield the minimum total loss $P_t$ listed in Eq. (22):

$$\partial P_t/\partial \varsigma =0,i_{\mathit{mq}}{i}_{\mathit{md}}=\text{constant}$$E23

Through derivation for Eq. (23) using Eq. (22), one can obtain:

$${\beta }_o={tan}^{-1}\sqrt{\frac{{\omega }_e^2{L}_d^2\left(R_s+R_c\right)+R_s{R}_c^2}{{\omega }_e^2{L}_q^2\left(R_s+R_c\right)+R_s{R}_c^2}}$$E24

Comments: (i) from Eq. (24) one can find that by neglecting the core loss ($R_c=\infty$), the commutation angle will become $\beta ={\beta }_{o1}={45}^{\circ }$; (ii) normally, $\beta >\left({\beta }_{o1}={45}^{\circ }\right)$ for pursuing the optimal efficiencies under varied operating conditions; (iii) as the commutation angle equation of Eq. (24) with nominal parameters of (${\overline{L}}_d,{\overline{L}}_q$, ${\overline{R}}_c$) is applied, the $\left(\beta ={\beta }_o\right)>{45}^{\circ }$ is resulted to yield better efficiency. If the accurate fitted parameters (${\widehat{L}}_d,{\widehat{L}}_q$, ${\widehat{R}}_c$) are available to determine the value of $\left(\beta ={\beta }_o\right)$, a slightly increased efficiency may further be obtained.

4.3 Measurement of motor key parameters

4.3.1 Winding resistance and inductances

The estimations of these parameters are similar to those of PMSM described previously.

4.3.2 Estimation of iron core loss resistance $R_c$

1. As shown in Figure 16, the SynRM is driven under various speeds at no-load with the commutation shift angle of $\beta =45°$. The input power $P_{\mathit{\text{in}}}$ and motor line currents are measured. Then the copper loss $P_{\mathit{cu}}$ can be calculated from Eq. (21), and the iron loss $P_c$ can be obtained as $P_c=P_{\mathit{\text{in}}}-P_{\mathit{cu}}$.

2. The d-axis and q-axis voltages across $R_c$ can be found from Figure 15a as:

   $$V_{\mathit{cd}}=V_{\mathit{ds}}-I_{\mathit{ds}}{R}_s,V_{\mathit{cq}}=V_{\mathit{qs}}-I_{\mathit{qs}}{R}_s$$E25

3. The equivalent resistance $R_c$ can be obtained from Eq. (21) and Figure 15a:

   $$P_c=\frac{3}{2}{R}_c\left({\left(\frac{V_{\mathit{cd}}}{R_c}\right)}^2+{\left(\frac{V_{\mathit{cq}}}{R_c}\right)}^2\right)\Rightarrow R_c=\frac{3}{2}\frac{\left(V_{\mathit{cd}}^2+V_{\mathit{cq}}^2\right)}{P_c}$$E26

Figure 16.

Core loss resistance estimation mechanism.

The estimated $R_c$ are ${\omega }_r=500\mathrm{rpm},R_c=34.71\mathrm{\Omega }\mathrm{;}{\omega }_r=1000\mathrm{rpm},R_c=38.84\mathrm{\Omega }\mathrm{;}{\omega }_r=1500\mathrm{rpm},$ $R_c=54.34\mathrm{\Omega }\mathrm{;}{\omega }_r=2000\mathrm{rpm},R_c=64.96$.

4.3.3 D-axis indexing

Taking an available three-phase SynRM (4-pole, 550 V, 2000 rpm, 3.7 kW, 17.67 Nm) as a test example, which is mechanically coupled with a three-phase PMSM (5 kW, 2000 rpm, 8-pole) as its dynamic load.

To verify the correctness of the detected rotor position, the AC voltage is injected into a-winding and b-winding terminals as indicated in Figure 14b. Owing to rotor saliency, a-phase current will be an amplitude modulated waveform. Let the PMSM be rotated at 100 rpm, the measured a-phase current $i_{\mathit{as}}$, Hall signal $H_a$, and rotor position ${\theta }_r$ due to the excited AC voltage 40 V/60 Hz which are plotted in Figure 17. It can be expected that the maximum and minimum current $i_{\mathit{as}}$ occur at $L_{\mathit{\text{asbs}},min}$ and $L_{\mathit{\text{asbs}},max}$, respectively. From Eq. (10), the $L_{\mathit{\text{asbs}},min}$ occurs at ${\theta }_r=\pi /3$, $L_{\mathit{\text{asbs}},max}$ occurs at ${\theta }_r=-\pi /6$, and the $i_{\mathit{as},max}$ and $i_{\mathit{as},min}$will also occur at ${\theta }_r=\pi /3$ and ${\theta }_r=-\pi /6$, respectively. Hence, from the measured waveforms in Figure 17, one can conclude that the detected rotor position is correct.

Figure 17.

Measured ($H_a,i_{\mathit{as}},{\theta }_r$) of an example of SynRM under excited AC voltage 40 V/60 Hz.

4.3.4 Back-EMF

From Eqs. (1), (3) and (15), one can find that SynRM possesses no back-EMF under no-load. As for the measurement mechanism proposed in Figure 18, a constant current is injected into the terminals c and b, and the a-phase terminal voltage $v_{\mathit{as}}\approx v_{\mathit{\text{an}}}^{\text{'}}$ is detected and shown in Figure 18, which can be expressed using Eqs. (1) and (3) with $i_{\mathit{bs}}=-i_{\mathit{cs}}=I$ and $i_{\mathit{as}}=0$ as:

Figure 18.

Configuration of current-injected back-EMF measurement arrangement and the measured Hall signal, back-EMF, and rotor position.

$$v_{\mathit{as}}=\frac{d}{\mathit{dt}}\left(\sqrt{3}{\mathit{IL}}_{B}sin2{\theta }_r\right)=2\sqrt{3}{\omega }_e{\mathit{IL}}_{B}cos2{\omega }_{e}t\underset{¯}{\underset{¯}{\mathrm{\Delta }}}Kcos2{\theta }_r,{\theta }_r={\omega }_{e}t$$E27

From the measured waveforms shown in Figure 18, one can be aware of some facts: (i) the adequacy of position sensing can be observed and (ii) the slotting effects can be comprehended from the ripples contaminated on $v_{a{n}^{\prime }}$.

4.4 Performance test of motor drive

Figure 19a and b shows the motor drive running characteristic test environment using eddy current brake and the suggested alternative test environment using load SPMSG as a dynamic load. Since the accurate eddy current brake and torque meter are not available, this alternative of loading test is worthy of adopting. However, the surface-mounted permanent-magnet synchronous generator (SPMSG) must be properly set, and it should be noted that the motor efficiency is both speed and load dependent.

Figure 19.

Test facilities for AC motor drive: (a) running characteristic test using eddy current brake and (b) running characteristic test using load SPMSG as dynamic load.

5. A battery/supercapacitor-powered EV PMSM drive

5.1 System configuration

Figure 20a shows the developed battery/super-capacitor (SC)-powered electric vehicle (EV) IPMSM drive with proper interface power converters. The control scheme motor drive is shown in Figure 20b, whereas the control schemes of other power stages are neglected here.

Figure 20.

The developed EV IPMSM drive: (a) schematic and (b) IPMSM drive control scheme.

5.1.1 IPMSM drive

5.1.1.1 System components

The used IPMSM is rated as three-phase 6-pole, 3000 rpm, 1 kW, 4.8 A, and 3.23 Nm. A surface-mounted PMSG is used as the dynamic load. The equipped dynamic brake leg is formed by ($T_{B1}$,$R_B$,$D_{B1}$) with the brake resistor $R_B=25\mathrm{\Omega }/200\mathrm{W}$.

5.1.1.2 Operation control

1. Motoring mode: the positive torque current command $i_{\mathit{qs}}^{\ast }$ yielded from the outer speed loop together with the properly set field excitation current command $i_{\mathit{ds}}^{\ast }$ let the IPMSM to yield positive developed torque for driving control. The improved generated torque can be obtained by commutation advanced shift and voltage boosting. The d-axis current command is set to yield the maximum developed torque within rated speed:

   $$i_{\mathit{ds}}^{\ast }=\frac{{\lambda }_m^{\prime }}{2\left(L_q-L_d\right)}-\sqrt{\frac{{\left({\lambda }_m^{\prime }\right)}^2}{4{\left(L_q-L_d\right)}^2}+{\left(i_{\mathit{qs}}^{\ast }\right)}^2}$$E28

2. Regenerative braking: as the speed command is decreased, the negative speed tracking error ${\epsilon }_{\omega }$ will let the torque current command $i_{\mathit{qs}}^{\ast }$ become negative automatically. The recovered motor power will be sent back to charge the battery.

5.1.2 Battery interface converter

5.1.2.1 System components

The battery bank is formed by two serially connected cells (UC BATTERY PS 40138, 72 V 30 Ah) with nominal voltage $V_b=156\mathrm{V}$. The DC-link voltage is arranged as $V_{\mathit{dc},min}=100\mathrm{V}$ $\left(\text{buck mode}\right),$ $V_{\mathit{dc},max}=400\mathrm{V}\left(\text{boost mode}\right)$.

5.1.2.2 Operation control

5.1.2.2.1 Discharging mode

As shown in Figure 20a, through the H-bridge DC/DC converter, the battery ($v_b=156\mathrm{V}$) establishes the adjustable and well-regulated DC-link voltage ($v_{\mathit{dc}}<v_b$ or $v_{\mathit{dc}}\ge v_b$). Here, the lowest and highest voltages are set as 100 V and 400 V, respectively. The buck converter is formed by ($T_{A+}$, $D_{A-}$, $L_B$, $D_{B+}$), while the boost converter is constructed using ($T_{A+}$, $T_{B+}$, $D_{B-}$, $L_{\mathrm{B}}$).

5.1.2.2.2 Charging mode

In making battery charging during motor regenerative braking, the DC-link voltage is regulated at the set value, and the front-end converter is operated in reverse direction automatically. The components ($T_{B-}$, $D_{B+}$, $L_B$, $D_{A+}$) and ($T_{A-}$, $D_{A+}$, $L_B$, $T_{B+}$) form the buck and boost converters, respectively. The maximum battery charging current is equivalently set by $i_{\mathit{Lm}}^{\ast }$ = 12 A of the current limiter.

5.1.3 SC interface converter

5.1.3.1 System components

The SC bank is rated as 6F/160 V, BMOD0006 E160 B02 (Maxwell Technologies), and maximum current = 170 V, maximum continuous current = 7 A, ESR (DC) = $240\mathrm{m\Omega }$. The system voltage and power ratings are set as $V_{\mathit{sc}}=100\mathrm{V}$, $V_{\mathit{dc}}=400\mathrm{V}$, and $P_{\mathit{dc}}=1\mathrm{kW}$.

5.1.3.2 Operation control

5.1.3.2.1 Motoring mode

In motoring mode, the DC-link voltage command $v_{\mathit{dc}}^{\ast }$ of the SC is set slightly higher (405 V) than the nominal DC-link voltage $V_{\mathit{dc}}\left(400\mathrm{V}\right)$. This will allow the switch $T_{C-}$ be operated to discharge the SC stored energy for acceleration. Then the battery is discharged subsequently as the energy of SC is exhausted.

5.1.3.2.2 Regenerative braking mode

As the braking is commanded, the DC-link voltage will be larger than its command due to the recovered motor stored kinetic energy. Hence the negative value of voltage tracking error will let the current command of current loop in the SC energy storage system become negative. The switch $T_{C+}$ will be operated to recycle the energy into the SC. Similarly, the maximum SC charging current is set by $20\mathrm{A}$ of the current limiter.

5.1.4 Adjustable DC-link voltage

For an inverter-fed motor drive, the lower DC-link voltage under lower speed and lighter load may yield smaller switching losses. Moreover, the EV is not often operated at relatively higher speed. As a result, the DC-link voltage can be properly set according to the motor running velocity to obtain higher efficiency over wide speed range. Most existing EV PCUs can only provide boosted motor drive DC-link voltage from battery. For the developed EV IPMSM drive, thanks to the flexibility possessed by the H-bridge DC/DC converter, the DC-link voltage can be lower than the battery voltage. The proposed DC-link voltage versus speed profile is depicted in Figure 21. The DC-link voltage profiles set for comparison are:

1. Fixed DC-link voltage: $v_{\mathit{dc}}=v_{\mathit{dc}1}=400\mathrm{V}$, $0\le {\omega }_r\le 4600\mathrm{rpm}$ ($0\le {\omega }_r\le {\omega }_m$)

2. Varied DC-link voltage:

   1. $v_{\mathit{dc}}=v_{\mathit{dc}2}=100\mathrm{V}$, $0\le {\omega }_r\le 1000\mathrm{rpm}$ ($0\le {\omega }_r\le {\omega }_1$).

   2. $100\mathrm{V}<\left(v_{\mathit{dc}}=v_{\mathit{dc}2}\right)\le 400\mathrm{V},$ $1000\mathrm{rpm}\le {\omega }_r\le 4600\mathrm{rpm}$ (${\omega }_1<{\omega }_r\le {\omega }_m$)

Figure 21.

The proposed DC-link voltage profile.

5.2 Some experimental results

5.2.1 Motor drive without SC

5.2.1.1 Steady-state characteristics

By setting $i_{\mathit{ds}}^{\ast }$ according to Eq. (26), Figure 22 shows the measured $H_a$ and ($i_{\mathit{as}}^{\ast }$, $i_{\mathit{as}}$, $i_{\mathit{ds}}^{\ast }$, $i_{\mathit{ds}}^{\prime }$) at ($v_b=156\mathrm{V}$, $v_{\mathit{dc}}=400\mathrm{V}$,$R_L=44.7\mathrm{\Omega }$, ${\omega }_r^{\ast }=3000\mathrm{rpm}$) of the developed EV IPMSM drive powered by battery H-bridge bidirectional DC/DC converter. The results show that the phase winding current commands $i_{\mathit{as}}^{\ast }$ are properly shifted with respect to the sensed Hall signal $H_a$, and the $i_{\mathit{as}}^{\ast }$ closely follows its command.

Figure 22.

Measured ${\mathit{H}}_{\mathit{a}}$ and (${\mathit{i}}_{\mathit{as}}^{\ast }$, ${\mathit{i}}_{\mathit{as}}$, ${\mathit{i}}_{\mathit{ds}}^{\ast }$, ${\mathit{i}}_{\mathit{ds}}^{\prime }$) of the developed standard IPMSM drive powered by H-bridge DC/DC front-end converter at $\left({\mathit{v}}_{\mathit{b}}=\mathit{156}V,{\mathit{\omega }}_{\mathit{r}}^{\ast }=\mathit{3000}\mathit{rpm},{\mathit{R}}_{\mathit{L}}=\mathit{44.7}\Omega \right)$ with ${\mathit{v}}_{\mathit{dc}}=\mathit{400}V$ and ${\mathit{i}}_{\mathit{ds}}^{\ast }=-\mathit{1}A$ being set (${\widehat{\mathit{i}}}_{\mathit{as}}=\mathit{5.15}A$).

5.2.1.2 Acceleration/deceleration/reversible and regeneration braking operation

Figure 23 shows the measured (${\omega }_r^{\ast }$, ${{\omega }^{\prime }}_r$) and ($i_{qs}^{*},{i^{\prime }}_{qs}$) at ($v_b=156\text{V},v_{dc}=400\text{V},R_L=100.7\Omega$) of the EV IPMSM drive powered by H-bridge DC/DC front-end converter under forward and reversible operations, with the speed command being set as (3000 rpm) to (−3000 rpm) and the rate of 600 rpm/s. In order to let the IPMSM drive be reversibly operated, the torque current component is made opposite automatically, which can be seen from the results. The smooth speed forward and reversible operations can be seen from the results.

Figure 23.

Measured (${\mathit{\omega }}_{\mathit{r}}^{\ast }$, ${\mathit{\omega }}_{\mathit{r}}^{\prime }$, ${\mathit{i}}_{\mathit{qs}}^{\ast }$, ${\mathit{i}}_{\mathit{qs}}^{\prime }$) of the developed IPMSM drive powered by H-bridge DC/DC front-end converter due to speed command change of (${\mathit{\omega }}_{\mathit{r}}=\mathit{0}\mathit{rpm}\to \mathit{3000}\mathit{rpm}\to \mathit{0}\mathit{rpm}\to -\mathit{3000}\mathit{rpm}\to \mathit{0}\mathit{rpm}$) under (${\mathit{v}}_{\mathit{b}}=\mathit{156}\mathit{V},{\mathit{v}}_{\mathit{dc}}=\mathit{400}\mathit{V},{\mathit{R}}_{\mathit{L}}=\mathit{100.7}\mathit{\Omega }$).

5.2.1.3 Comparative evaluation for fixed and varied-voltage DC-links

The measured ($v_{\mathit{dc}}^{\ast }$, $v_{\mathit{dc}}^{\prime }$, ${\omega }_r^{\ast }$, ${\omega }_r^{\prime }$, $v_b$, $i_b$, $P_b$, $E_b$) of the EV IPMSM drive powered by H-bridge DC/DC converter at ($R_L=75\mathrm{\Omega }$) under the speed pattern defined by the ECE15. The speed pattern defined by ECE15 with fixed $v_{\mathit{dc}}=$400 V and varied $v_{\mathit{dc}}$ are compared in Figure 24a and b, where $P_b\equiv v_b\times i_b$ and $E_b\equiv \int P_b\mathit{dt}.$ The developed EV IPMSM drives with fixed and varied $v_{\mathit{dc}}$ all yield satisfactory driving characteristic.

Figure 24.

Measured (${\mathit{v}}_{\mathit{dc}}^{\ast }$, ${\mathit{v}}_{\mathit{dc}}^{\prime }$, ${\mathit{\omega }}_{\mathit{r}}^{\ast }$, ${\mathit{\omega }}_{\mathit{r}}^{\prime }$, ${\mathit{v}}_{\mathit{b}}$,${\mathit{i}}_{\mathit{b}}$, ${\mathit{P}}_{\mathit{b}}$, ${\mathit{E}}_{\mathit{b}}$) of the developed EV IPMSM drive powered by H-bridge DC/DC front-end converter at (${\mathit{R}}_{\mathit{L}}=\mathit{75}\mathit{\Omega }$) due to the speed pattern defined by the ECE15: (a) ${\mathit{v}}_{\mathit{dc}}$ is fixed at $\mathit{400}\mathit{V}$ and (b) ${\mathit{v}}_{\mathit{dc}}$ is varied with speed.

The measured $E_b$ by the fixed $v_{\mathit{dc}}$ and the varied $v_{\mathit{dc}}$ corresponding to Figure 24 are further compared in Figure 25. From the comparison one can find that the adjustable DC-link voltage can reduce battery energy consumption.

Figure 25.

Measured ${\mathit{E}}_{\mathit{b}}$ of the developed standard IPMSM drive powered by H-bridge converter with fixed ${\mathit{v}}_{\mathit{dc}}=\mathit{400} \mathit{V}$ and varied ${\mathit{v}}_{\mathit{dc}}$ due to speed pattern defined by the ECE15 at (${\mathit{R}}_{\mathit{L}}=\mathit{75}\mathit{\Omega }$).

5.2.2 Motor drive incorporated with SC

Under the preset speed changing pattern as shown in Figure 26, the measured results of the established EV IPMSM drive powered by H-bridge DC/DC front-end converter with acceleration and deceleration rates of 750 rpm/s at ($v_b=156\mathrm{V},v_{\mathit{dc}}^{\ast }=400\mathrm{V},v_{\mathit{sc}}^{\ast }=405\mathrm{V},R_L=75\mathrm{\Omega }$) are provided as: Figure 26a: (${\omega }_r^{\ast }$, ${\omega }_r^{\prime }$, $i_b$, $v_b$, $v_{\mathit{dc}}$) without SC; Figure 26b: (${\omega }_r^{\ast }$, ${\omega }_r^{\prime }$, $i_b$, $i_{\mathit{sc}}$, $v_{\mathit{sc}}$, $v_b$, $v_{\mathit{dc}}$) with SC. Without SC, the battery possesses larger charging and discharging currents as shown in Figure 26a.

Figure 26.

Measured results of the established standard EV IPMSM drive powered by H-bridge DC/DC front-end converter due to preset speed pattern with acceleration and deceleration rates of 750 rpm/s at (${\mathit{v}}_{\mathit{b}}=\mathit{156}\mathit{V},{\mathit{v}}_{\mathit{dc}}^{\ast }=\mathit{400}\mathit{V},{\mathit{v}}_{\mathit{sc}}^{\ast }=\mathit{405}\mathit{V},{\mathit{R}}_{\mathit{L}}=\mathit{75}\mathit{\Omega }$): (a) ((${\mathit{\omega }}_{\mathit{r}}^{\ast }$, ${\mathit{\omega }}_{\mathit{r}}^{\prime }$), ${\mathit{i}}_{\mathit{b}}$, ${\mathit{v}}_{\mathit{b}}$, ${\mathit{v}}_{\mathit{dc}}$) without SC and (b) ((${\mathit{\omega }}_{\mathit{r}}^{\ast }$, ${\mathit{\omega }}_{\mathit{r}}^{\prime }$), ${\mathit{i}}_{\mathit{b}}$, ${\mathit{i}}_{\mathit{sc}}$, ${\mathit{v}}_{\mathit{sc}}$, ${\mathit{v}}_{\mathit{b}}$, ${\mathit{v}}_{\mathit{dc}}$) with SC under discharging operation.

The application of SC can reduce the battery currents in driving and regenerative braking operations under the same scenario. In the results presented in Figure 26b with SC using active discharging approach (noted that $v_{\mathit{sc}}^{\ast }>v_{\mathit{dc}}^{\ast }$), the battery current remains zero at the beginning under driving mode, the SC discharges via its bidirectional converter first, and the battery pack does not supply power to DC link until the SC is unable to maintain the driving torque. Hence, not only the energy of regenerative braking can be recycled, but also the battery can avoid its intermittent charging/discharging operations. Obviously, by adding the supercapacitor storage system, the battery provides the fewer power during acceleration condition, and its current would never become negative. Besides, the battery voltage is also less fluctuated.

6. A bidirectional SMR-fed synchronous reluctance motor drive

6.1 System configuration

Figure 27a shows the developed SMR-fed SynRM drive. And the control scheme of the motor drive is depicted in Figure 27b. The three-phase full-bridge SMR is employed to power the motor drive from the mains with good line drawn power quality. The boosted and well-regulated DC-link voltage enhances motor driving performance over wide speed range. Moreover, the recovered energy during regenerative braking can be successfully sent back to utility grid.

Figure 27.

The SMR-fed SynRM drive: (a) schematic and (b) motor drive control scheme.

In treating the driving control for a SynRM drive, some distinct features are worth noting: (i) since only reluctance torque is exited, the proper commutation angle setting is the most critical issue affecting its developed torque characteristics; (ii) the iron loss of SynRM is significant and varied with operating conditions; and (iii) the d-axis inductance is significantly changed with the armature current level due to magnetic saturation; hence, the iron loss and copper loss should be simultaneously considered to yield the improved overall efficiency. In the SynRM drive control scheme shown in Figure 27b, the proposed improved approaches include (i) the commutation angle is set by an adaptive commutation scheme (ACS) using online estimated d-axis inductance to minimize motor total loss. Under higher speed, an extra commutation angle is yielded by the field-weakening commutation scheme (FWCS) to limit the voltage. And (ii) in the proposed ramp-comparison current-controlled PWM (RC-CCPWM) scheme, the basic feedback control is augmented with a back-EMF elimination feedforward controller and a transient cross-coupling field controller to enhance the current control performance.

6.1.1 SynRM drive

6.1.1.1 System components

1. The employed SynRM is rated as SynRM: 4-pole, 550 V, 2000 rpm, 3.7 kW. The inverter is formed using three Mitsubishi IGBT modules CM100DY-12H ($V_{\mathit{CE}}=600\mathrm{V},I_C=100\mathrm{A},$ $I_{\mathit{CM}}=200\mathrm{A}$). The SPWM switching is employed. A PMSG with load resistance$R_L$ is served as the dynamic load of the studied SynRM.

2. Estimated motor parameters:$R_s=0.47\mathrm{\Omega }$, $R_c=61.37\mathrm{\Omega },L_q=28.92\mathrm{mH}$ (at 66.6 Hz). The d-axis inductance $L_d$ is much affected by the magnetic saturation compared to the q-axis inductance $L_q$. The fitted ${\widehat{L}}_d\left(i_{\mathit{md}}\right)$ (in H) is expressed as:

$${\widehat{L}}_d\left(i_{\mathit{md}}\right)=\left\{\begin{array}{c}0.0559\left(\mathrm{H}\right),i_{\mathit{md}}\le 3\mathrm{A}\\ {\alpha }_{1}ln\left(i_{\mathit{md}}\right)+{\alpha }_2\left(\mathrm{H}\right),i_{\mathit{md}}>3\mathrm{A}\end{array}\right.$$E29

where ${\alpha }_1=-0.00864$ and ${\alpha }_2=-0.06458.$

6.1.1.2 Commutation scheme

1. The proposed adaptive commutation scheme (ACS): the fitted ${\widehat{L}}_d\left(i_{\mathit{md}}\right)$ and the estimated motor parameters are applied to Eq. (24) to determine the commutation angle $\left(\beta ={\beta }_o\right).$

2. Field-weakening commutation scheme (FWCS)

Under higher speeds, the field weakening via commutation advanced shift must be applied to satisfy the electrical capabilities of a SynRM drive expressed as:

$${\widehat{I}}_s=\sqrt{i_{\mathit{qs}}^2+i_{\mathit{ds}}^2}\le {\widehat{I}}_{\mathit{sm}}$$E30

$${\widehat{V}}_s=\sqrt{v_{\mathit{qs}}^2+v_{\mathit{ds}}^2}\le {\widehat{V}}_{\mathit{sm}}=v_{\mathit{dc}}/2$$E31

where ${\widehat{I}}_s\left({\widehat{V}}_s\right)$ = phase current (voltage) magnitude and the maximum voltage ${\widehat{V}}_{\mathit{sm}}=v_{\mathit{dc}}/2$ is set for a SPWM inverter.

For regulating the voltage limit automatically, the commutation shift angle ${\beta }_f$ is yielded by the proposed FWCS scheme shown in Figure 27b as:

$${\beta }_f=\left\{\begin{array}{c}0,\sqrt{v_{\mathit{qs}}^{\ast 2}+v_{\mathit{ds}}^{\ast 2}}<\left(\frac{v_{\mathit{dc}}^{\prime }}{2}\right)\\ G_{\mathit{cf}}\left(s\right)\left(\sqrt{v_{\mathit{qs}}^{\ast 2}+v_{\mathit{ds}}^{\ast 2}}-\left(\frac{v_{\mathit{dc}}^{\prime }}{2}\right)\right),\sqrt{v_{\mathit{qs}}^{\ast 2}+v_{\mathit{ds}}^{\ast 2}}\ge \left(\frac{v_{\mathit{dc}}^{\prime }}{2}\right)\end{array}\right.$$E32

6.1.2 Three-phase full-bridge SMR

To simultaneously preserve bidirectional power flow and power factor corrected capabilities, the three-phase full-bridge SMR is established and employed as the AC front end of the SynRM drive. The three-phase AC input line voltage is 220 V/60 Hz. And the DC-link voltage is set as $V_{\mathit{dc}}$ = 550 V. Three IGBT modules CM100DY-12H are used to form the power circuit.

6.2 Some experimental results

6.2.1 Performance of SMR front end

Figure 28a and b shows the steady-state characteristics of the three-phase full-bridge SMR-fed SynRM drive at ($V_{\mathit{dc}}=550\mathrm{V}$,${\omega }_r=2000\mathrm{rpm}$, $\beta ={\beta }_{o1}=45°$, $R_L=13.7\mathrm{\Omega }$). The results indicate that the SMR possesses good operating characteristics in powering the followed SynRM drive, good line drawn power quality, and well-regulated DC-link voltage which are seen.

Figure 28.

Measured steady-state results of the developed SMR-fed SynRM drive under (${\mathit{V}}_{\mathit{dc}}=\mathit{550}\mathit{V},{\mathit{\omega }}_{\mathit{r}}^{\prime }=\mathit{2000}\mathit{rpm},\mathit{\beta }=\mathit{45}°,{\mathit{R}}_{\mathit{L}}=\mathit{13.7}\mathit{\Omega }$): (a) (${\mathit{v}}_{\mathit{dc}},{\mathit{v}}_{\mathit{A}},{\mathit{i}}_{\mathit{A}}$) and (b) (${\mathit{i}}_{\mathit{A}},{\mathit{i}}_{\mathit{B}},{\mathit{i}}_{\mathit{C}}$).

6.2.2 Effectiveness of the proposed ACS scheme

Under ($V_{\mathit{dc}}=550\mathrm{V}$,$R_L=13.7\mathrm{\Omega }$), let the commutation angles be $\beta ={\beta }_{o1}={45}^{\circ }$ and $\beta ={\beta }_o$ yielded from Eq. (24); the measured (${\omega }_r^{\prime }$, ${\widehat{I}}_s$, $P_{\mathit{dc}}$) of the SynRM drive at two speeds are compared in Figure 29a and b. One can find that the proposed commutation angle setting can lead to less power consumption and the reduction of current magnitude.

Figure 29.

Measured (${\omega }_r^{\prime }$, ${\widehat{I}}_s$, $P_{\mathit{dc}}$) of the established SMR-fed SynRM drive with $\beta ={\beta }_{o1}={45}^{\circ }$ and $\beta ={\beta }_o$ by the proposed ACS under ($V_{\mathit{dc}}=550V$,$R_L=13.7\Omega$): (a) ${\omega }_r^{\prime }=1000\mathit{rpm}$ and (b) ${\omega }_r^{\prime }=2000\mathit{rpm}$.

6.2.3 Effectiveness of the proposed FWCS in high-speed driving performance

Since the rated speed of the used SynRM is 2000 rpm, the proper field weakening is necessary under the speeds larger than 2000 rpm. Let the commutation angle be $\beta ={\beta }_{o1}+{\beta }_f$ with ${\beta }_{o1}={45}^{\circ }$ being set and ${\beta }_f$ being yielded by the developed FWCS. Figure 30 plots the measured (${\omega }_r^{\ast },{\omega }^{\prime }{}_r$) and the yielded commutation angle ${\beta }_f$ during starting process at ($V_{\mathit{dc}}=550\mathrm{V}$,$R_L=303.9\mathrm{\Omega }$) from 0 to 4000 rpm with an acceleration rate of ($250\mathrm{rpm}/\mathrm{s}$). One can find from the results that speed tracking error under high speeds can be eliminated by applying the FWCS via increasing the commutation angle component ${\beta }_f$.

Figure 30.

Measured (${\omega }_r^{\ast },{\omega }^{\prime }{}_r$) and the commutation angle ${\beta }_f$ yielded by the FWCS during starting process from 0 to 4000 rpm at ($V_{\mathit{dc}}=550\mathrm{V}$,$R_L=303.9\mathrm{\Omega }$).

6.2.4 Reversible operation

The measured (${\omega }_r^{\ast },{\omega }_r^{\prime }$) and ${\widehat{I}}_s$ of the SMR-fed SynRM drive at ($V_{\mathit{dc}}=550\mathrm{V}$,$R_L=13.7\mathrm{\Omega }$) with the commutation angle $\beta ={\beta }_o$ due to the ramp speed command change ${\omega }_r^{\ast }=0\to 2000\mathrm{rpm}\to 0\to -2000\mathrm{rpm}\to 0$ with the rising and falling rates of 250 rpm/s being plotted in Figure 31. The results indicate the smooth speed tracking and reversible operation of the developed SynRM drive.

Figure 31.

Measured (${\mathit{\omega }}_{\mathit{r}}^{\ast },{\mathit{\omega }}_{\mathit{r}}^{\prime }$) and ${\widehat{\mathit{I}}}_{\mathit{s}}$ of the SynRM drive at (${\mathit{V}}_{\mathit{dc}}=\mathit{550}\mathit{V}$,${\mathit{R}}_{\mathit{L}}=\mathit{13.7}\mathit{\Omega }$) with $\mathit{\beta }={\mathit{\beta }}_{\mathit{o}}$ due to ramp speed command change ${\mathit{\omega }}_{\mathit{r}}^{\ast }=\mathit{0}\to \mathit{2000}\mathit{rpm}\to \mathit{0}\to -\mathit{2000}\mathit{rpm}\to \mathit{0}$ with both the rising rate and falling rate being 250 rpm/s.

6.2.5 Regenerative braking

The SynRM drive is first stably operated at (${\omega }_r^{\prime }=2000\mathrm{rpm},V_{\mathit{dc}}=550\mathrm{V},R_L=\infty$). Now the speed command is set from ${\omega }_r^{\ast }=$ 2000 rpm to 0 rpm linearly in 1 s. The measured DC-link voltage $v_{\mathit{dc}}$(${\omega }_r^{\ast },{\omega }_r^{\prime }$), ${\widehat{I}}_s$ and phase-a winding current are of the established SMR-fed SynRM drive which are plotted in Figure 32a. One can be aware of the results that the current command ${\widehat{I}}_s$ becomes negative under regenerative braking operation. Successful SynRG operation during regenerative braking is confirmed. And Figure 32b shows the DC-link voltage $v_{\mathit{dc}}$, phase-A voltage $v_A$, and current $i_A$ of the mains. From Figure 32b, the regenerative braking of SynRM drive with energy recovered back to the mains can be observed.

Figure 32.

Measured results of the established SMR-fed SynRM drive during braking by letting the speed command be changed from ${\mathit{\omega }}_{\mathit{r}}^{\ast }=\mathit{2000}\mathit{rpm}$ to 0 rpm linearly in 1 s: (a) DC-link voltage ${\mathit{v}}_{\mathit{dc}}$, (${\mathit{\omega }}_{\mathit{r}}^{\ast },{\mathit{\omega }}_{\mathit{r}}^{\prime }$), current command ${\widehat{\mathit{I}}}_{\mathit{s}}$, and a-phase winding current, ${\mathit{i}}_{\mathit{as}}$ and (b) DC-link voltage ${\mathit{v}}_{\mathit{dc}}$, A-phase voltage, and current (${\mathit{v}}_{\mathit{A}},{\mathit{i}}_{\mathit{A}}$) from the mains.

6.2.6 Efficiency evaluation

In order to verify the efficiency enhancement of the established SMR-fed SynRM drive using the proposed ACS, the measured steady-state characteristics at ($V_{\mathit{dc}}=550\mathrm{V}$, $R_L=13.7\mathrm{\Omega }$,${\omega }_r^{\prime }=2000\mathrm{rpm}$) with the commutation angles $\beta ={\beta }_{o1}={45}^{\circ }$ and $\beta ={\beta }_o$ yielded by the proposed ACS are listed in Table 1. The efficiencies are defined as:

Speed	$\mathit{\beta }={\mathit{\beta }}_{\mathit{o}\mathit{1}}={45}^{\circ }$			$\mathit{\beta }={\mathit{\beta }}_{\mathit{o}}$ by ACS
2000 rpm	$P_L$ = 3253.8 W			$P_L$ = 3251.7 W
	$P_{\mathit{dc}}$ = 4064.3 W			$P_{\mathit{dc}}$ = 3980.1 W
	$P_{\mathit{ac}}$ = 4380.0 W			$P_{\mathit{ac}}$ = 4285.0 W
	$\mathit{PF}$ = 0.998	${\mathit{THD}}_{i_A}$ = 5.05%		$\mathit{PF}$ = 0.997	${\mathit{THD}}_{i_A}$ = 4.98%
	${\eta }_{\mathit{SMR}}$ = 92.8%	${\eta }_{\mathit{1}}$ = 80.1%	$\eta$ = 74.3%	${\eta }_{\mathit{SMR}}$ = 92.9%	${\eta }_{\mathit{1}}$ = 81.7%	$\eta$ = 75.9%

Table 1.

Measured steady-state characteristics of the established SMR-fed SynRM drive at ($V_{\mathit{dc}}=\mathit{550}\mathit{V},R_L=\mathit{13.7}\mathit{\Omega },{\omega }_r^{\prime }=\mathit{2000}\mathit{rpm}$).

${\eta }_{\mathit{SMR}}\equiv P_{\mathit{dc}}/P_{\mathit{ac}}=\mathrm{SMR}\text{efficiency}$, ${\eta }_m\equiv \text{SynRM efficiency}$, ${\eta }_g\equiv \text{load PMSG efficiency},$ ${\eta }_1\equiv P_L/P_{\mathit{dc}}=\left(P_m/P_{\mathit{dc}}\right)\left(P_g/P_m\right)\left(P_L/P_g\right)$, and $\eta \equiv {\eta }_{\mathit{SMR}}{\eta }_1=P_L/P_{\mathit{ac}}=\text{total efficiency}$.

The improvement in energy conversion efficiency via the proposed ACS is obvious from the results. The high efficiency possessed by the developed SynRM drive can also be aware.

7. Conclusions

As generally recognized, PMSM and SynRM are the two new motors having the potential to achieve the IE4-class and even the IE5-class efficiency. During the past years, the popularly used induction motors have been gradually replaced by these two motors in various applications, such as home appliances, EVs, wind generators, etc. For facilitating the development of high-performance PMSM and SynRM drives, this article has presented some basics and key affairs. Moreover, an example of EV PMSM drive and an example of SMR-fed SynRM drive have also been introduced for demonstration. Generally speaking, the most critical issues lie in (i) selecting suited type of motor for specific applications, (ii) proper commutation instant setting and shifting, (iii) applying DC-link voltage boosting approach to enhance high-speed driving characteristics, and (iv) properly conducting the regenerative braking operation control.